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    • 专利摘要:

    公开号:NL8500542A
    申请号:NL8500542
    申请日:1985-02-26
    公开日:2003-02-03
    发明作者:
    申请人:Secr Defence Brit；
    IPC主号:
    专利说明:

    ase-controlled reflector element.
    Field of technology
    The invention relates to a phase-controlled reflector element that can operate at microwave frequencies.
    Phase shifting reflector systems are useful for a wide range of applications. They find application in beam shaping and beam control - that is, used in combination with a transmitter, they can be useful for varying either the shape of the main beam and the side lobes or the direction of the main beam. This is achieved by selecting and varying the phase input by each system element. They can also be used in beam selection - that is, they can be used for direct radiation incident from one or more selected directions on a receiver. They also find application with signal modulation. The phase introduced by each reflector element can be varied coherently in a time-dependent manner for obtaining frequency modulation. Reflector elements capable of independent polarization control can also be used in combination with an analyzer for effecting amplitude modulation or signal selection through gate action.
    Background of the prior art
    A known phase-shift system for frequencies in the range of 3 to 8 GHz comprises a system of horn-fed receiver antennas arranged back-to-back with the same system of transmitter antennas, each with a horn output. Corresponding receive and transmit antennas are coupled in pairs via respective phase shift networks. This typical transmission system is expensive, takes up a lot of space and is very heavy. It would have a volume in the range of, for example, 1 m · *.
    There is currently a need for phase shifting systems that can operate at high frequencies, in particular at microwave frequencies in the range of 3 to 100 GHz. A prior art system is a very unattractive option because of its price and volume.
    Description of the invention
    The invention has for its object to provide phase control elements which are robust, have a low weight, are compact and can be manufactured relatively inexpensively. These elements and systems are intended for microwave radiation in the frequency range of 3 to 100 GHz.
    The invention therefore provides a phase control element comprising: (1) a dipole, (2) a substantially lossless dielectric member disposed near the dipole and designed such that radiation is strongly coupled thereto, and (3) a variable reactance designed as a substantially lossless load of the dipole, whereby the radiation falling on the dipole is retransmitted with a phase variable in accordance with the sign and the strength of the load reaction.
    The material of the dielectric member is selected such that it has low dielectric losses because the absorbed micro wave power is small compared to the power coupled to or from the dipole by the dielectric member. The designation "substantially lossless dielectric member" will be further explained.
    An additional resistance contribution to the load impedance results from the non-ideal properties of the load. A certain low resistance contribution is unavoidable. A requirement is that as much radiation as possible that falls on the dipole is reflected. Power absorbed by the load will be low and therefore the reflectivity will be high, provided that either the impedance of the load in strength is comparable to the impedance of the dipole and the resistance part of the load impedance is small compared to the reactive part or that the impedance of the load is either very high or very low in strength compared to the dipole impedance. In connection with this, it is noted that, in microwaves theory, open circuits and short circuits are usually treated as extremes of reactances; the expression "reactance", "reactive" and similar designations will be further elucidated accordingly to include, inter alia, open circuits and short circuits.
    It is of particular advantage that the dipole and its load can be constructed in planar form. The dielectric member may cover a volume of the order of 10 and the dipole and the load of -7 ', a combination three orders of magnitude smaller than the known devices. It is also advantageous that the dipole in the radiation mainly couples only on one side due to the strongly coupling dielectric member, which simplifies efficient adaptation to a microwave field.
    The phase control element can be of hybrid construction. The dipole can be formed from metal applied to the surface of a substrate of insulating dielectric material. The load would in this case comprise discrete components attached to form a network connected in parallel with the dipole.
    The phase-controlled element can be of integrated construction, i.e. the dipole can be provided with a substrate of substantially loss-free semiconductor material. The substrate can also be a composite body with a surface of such semiconductor material. In the case of the latter, the impedance components can be formed as components integrated with the semiconductor material. The substrate may also consist of insulating dielectric material and the phase control element in its construction may comprise a supporting layer of semiconductor material, the dipole being located between the dielectric member and this layer. In this modified embodiment, heat dissipation members can be used without many problems. The layer of semiconductor material can be supported by metal or by a thin layer of electrically insulating dielectric material with a metal coating. This modified embodiment is therefore preferable for high-power applications and in this case efficient heat dissipation is important.
    The invention uses the following principle. A variable reactance is connected in parallel with the dipole. This dipole shines with unchanged polarization, but with a phase shift that is given by a complex reflection power Ry:
    where G ^ + jB ^ is the dipole admittance as a radiation source and load admittance. Ry is the voltage reflection potential. It should be noted that Ry has the unit module as long as the load condition is zero. This ideal case is dependent on the loss-free impedance components and the absence of absorbed power in the dipole metal and the dielectric member. The phase shift of the re-radiated signal relative to the incident signal is in the general case:
    In the lossless case = 0, the phase shift becomes
    If there is a variable over a range from a strong negative to a strong positive value, a phase variation of almost -ft to 1t can be achieved. This degree of phase control requires a load that must be variable from inductive to capacitive.
    Where the phase control element comprises a single dipole, the element will only be coupled to radiation with a polarization component that runs parallel to the dipole. Power re-emitted by this dipole will in turn only be polarized in parallel with the dipole.
    The network may, for example, comprise several impedance components to be selected by switches, each component comprising the combination of a reactance and a control switch.
    As a further example, the phase control element may comprise a crossed pair of orthogonal dipoles, one dipole load being either an open circuit or a short circuit, while the other dipole load being an anti-parallel pair of diodes. In this construction the load impedance is dependent on the level of the incident radiation power. The load impedance is high at low levels. At high levels, however, the diodes start conducting and the load impedance becomes low.
    A more versatile embodiment of the invention comprises a crossed pair of orthogonal dipoles, each provided with independently controllable loads. In this construction, each dipole is formed and configured to serve as an inductive load that is connected in parallel with the others. This construction permits separate phase shifts for each of the two orthogonal polarizations - the polarization directions parallel to each dipole. Thus, if the incident polarization is a circular polarization (independent of the direction of rotation) or a flat polarization at an angle of 45 ° with respect to the dipoles, the choice of phase shifts for each dipole allows the polarization of the re-irradiation either is circularly polarized with one of both directions of rotation or flat with an angle of + - 45 ° - that is, the polarization change is also possible.
    Systems can be constructed using many equal, single or crossed dipoles. A common dielectric member can also be used.
    Brief description of the drawings with this description
    Figures 1 and 2 show, in top plan view and in cross-section, respectively, a phase-controlled reflector element with a single dipole according to the invention; 3 and 4 show, in top plan view and in cross-section, respectively, a phase control element with a crossed dipole; Figs. 5 and 6 each show flat cross-sections of the control element in the above-mentioned Figs. 3 and 4, and show more detailed different control circuit configurations; Fig. 7 shows a cross-section of an FM phase modulator with a phase control element in the form of a single crossed dipole; Fig. 8 shows a cross-section of a control device for directing a bundle included in a set of dipoles; 9 and 10 show in plan view two modified structures of a phase control element with a crossed dipole; Fig. 11 is a cross-sectional view of a duplex radar system provided with a set of crossed dipoles each as shown in one of the preceding Figs. 9 and 10; Fig. 12 shows a phase control element provided with a shortened transmission line and a reactive load in the form of a varactor diode; Fig. 13 shows a crossed dipole as a phase control element provided with varactor diodes; and Fig. 14 shows a cross-section of a transmitter with controllable directional effect.
    Description of embodiment
    Some embodiments of the invention will now be described with reference to the accompanying drawings, but only by way of example.
    Figures 1 and 2 show an example of a phase-controlled reflector element 1 with a single dipole according to the invention. This element comprises a single dipole 3 formed from metal applied to the substrate 5 of nearly zero loss-free dielectric material, for example silicon-semiconductor material. In this embodiment, the substrate 5 acts both as a dipole carrier and as a dielectric member for coupling radiation to the dipole 3. The dipole 3 is subdivided into two legs 3a, 3b of equal or substantially equal length. A local impedance network 7, located in the vicinity of the center of the dipole 3, is connected between the two legs 3a, 3b. This network 7 comprises a shortened transmission line 9 which serves as an inductive load. The network 7 also comprises a plurality of switch-selectable impedance components 11, 13, each of which in this example consists of a capacitor 11c, 13c and a PIN diode switch 1s, 13s. At appropriate values of self-induction and capacitance, the operation of the switches lis, 13s produces a net load over the dipoles 3 which can be either inductive or capacitive. Each of the capacitors 11c and 13c is or is not connected across dipole 3, depending on whether its associated diode switch 1s or 13s forms a short circuit or an open circuit, respectively. This provides four reactivity options selectable by a two-bit instruction. The control lines 15, 17, 19 serve for bias control. The control line 15 is common to the two diodes lis and 13s, while the lines 17 and 19 are connected to each of the diodes lis and 13s, respectively. Pre-voltages applied to the control lines 15 and 17, 15 and 19 switch the diodes lis and 13s, which in turn connect the capacitors 11c and 13c across the dipole 3. Parasitic coupling between the dipole 3 and the control lines 15, 17 and 19 is minimized by laying the lines in a direction perpendicular to the dipole 3.
    While the impedance network 7 comprises a fixed inductance with switchable capacitors, a switchable self-induction with a fixed capacitor can also be applied.
    Attention will now be paid to factors which determine the choice of the length of the dipole 3. In the case of resonance, the length "& V of the dipole and the absolute wavelength λν of the radiation are given by the formula
    (1) (See Brewitt-Taylor et al. "Planar Antennas on a dielectric surface", Electronics Letters Vol. 17, No. 20, pp. 729-731 (October 1981)). wherein έ j and ^ 2 are the dielectric constants of the media on either side of the dipole. For silicon 12 and air £ 2 Cf * *
    The symbol X represents the wavelength of the radiation measured in the dielectric substrate medium. This formula assumes the resonance according to the lowest mode - so-called "half wavelength" resonance, analogous to the resonance in a free-standing dipole. At this wavelength, the resonance of the next higher order corresponds to a length that is three times this value. The length of the dipole € is chosen within this area:
    (2)
    The formula (1) given above is theoretical in that it assumes a dipole length / width aspect ratio approaching infinitely. However, this formula can also be considered as a reasonable approximation for a dipole with an aspect ratio of 10: 1. The formula can be changed by a simple geometric factor to take into account the dipole shape and the aspect in more special cases.
    The damping losses due to the specific resistance of the mounting substrate or the dielectric member are approximated by the ratio (Z / p g) in which Z is the characteristic impedance and g is the specific blade resistance. For a silicon substrate (Z1 ^ -100 ohm) of nominal thickness of 400 µm, a specific resistance of 100 ohm.cm corresponds to a damping loss of approximately 5%, an acceptable value. The antenna dipole impedance and the polar radiation diagram are also sensitive to the specific substrate resistance, but for the dipole described, this effect is small for specific substrate resistances of 100 ohm.cm and higher.
    The shortened length of the transmission line 9 is typically between X / 32 and X / 8 and is therefore inductive. eff eff
    A more versatile variant of the above-mentioned control element 1 is shown in top view and in cross-section in Figs. 3 and 4. This element 1 comprises a pair of crossed orthogonal dipoles 3 and 3 'formed as a pattern from a common layer of metal applied to the surface of a thin layer 21 of semiconductor silicon - a layer 21 having a thickness of in particular between X / 100 and X / 4, wherein X is the selected signal wavelength measured in the silicon. A protective oxide layer 23 is provided between the metal and the silicon in order to prevent the formation of undesired intermetallic compounds. The silicon layer 21 is supported by a thin coating of berylium oxide 25 and a metal coating 27 to facilitate heat dissipation. The dipoles 3 and 3 "are mounted next to or just above the surface of a dielectric member 5 of insulating dielectric material. The dielectric constant of this insulating material 5 is chosen such that the dipoles essentially only link the radiation incident through the material 5.
    Each of the dipole legs 3a, 3b, 3'a, 3'b has a respective slot 4a, 4b, 4'a, 4'b. Each slotted dipole portion serves as a shortened transmission line such as 9, connected in parallel with a respective dipole leg 3a, 3b, 3'a or 3'b, each leg having an approximate length of 4. The shorter line length, i.e. the length of each slot, is less, especially in the range of> / 32 to λ / 8 and thus each shortened line offers an inductive load. These parallel inductive loads across the dipoles 3 and 3 'are implemented by using impulse components 11, 13 and 11', 13 'to be selected with the aid of switches. Each of the impedance components 11, 11 ', 13 and 13' to be selected by switches comprises a capacitor 11c, 11'c, 13c or 13'c and a PIN diode switch 11s, Us, 13s and 13s, respectively.
    The loaded dipoles 3 and 3 'link independently to their own polarization. The phase shifts introduced into the re-irradiated fields are driven by impedance components 11, 13, 11 'and 13' and are independent.
    Considering now, an incident radiation plane is polarized at 45 ° for the dipoles 3 and 3 ', which induce currents in phase. The re-radiated fields are subject to phase shifts and Θ for the horizontal and vertical dipoles 3 'and 3, respectively. If Θ = Cp, the resulting radiation is flatly polarized at 45 ° (i.e. parallel to the incident field). On the other hand, if 0 = ^ + 1¾ ^, the re-radiated field is flatly polarized at -45 ° (i.e., perpendicular to the incident field). If Θ = _ + "Ttï / 2, circular polarization of one of the rotational directions is re-irradiated. In any case, the re-radiated field is shifted in phase by φ relative to the incident field. This demonstrates the independent control of the phase and the polarization.
    The control line connection to the PIN diodes lis, 11s, 13s and 13s can be made via resistor layers. It is also possible to place low-frequency semiconductor devices under the antenna metal, for providing logic functions or for driving the PIN diodes lis, Us, 13s and 13s. Electric power can be supplied here either through further transmission lines or via resistor connections.
    If Strong microwave powers are to be controlled by the relay elements, the power supply required for the PIN diodes lis will be increased to 13s (in particular to about 10 mA for a diode capable of controlling 10 W microwave power). For the crossed dipoles 3, 3 ', it may be useful to supply the current for all control diodes due to the energy dissipation through resistor connections. One way to avoid this problem is to rectify a small amount of incident microwave power to provide the direct current for the diodes 1 to 13s and for all logic and control transistors present. Only control signals with a low level need then be supplied by the resistor connections. Schottky barrier diodes are suitable as HF-DS power converters. In the circuit shown in FIG. 5, a metal line 11m and two Schottky barrier rectifying diodes 11r are connected in series via a dipole slot 4'a. The diodes 11r are coupled to the microwave field via the line 11m and connected by a capacitor C at 10'a to the dipole leg 3'a. The rectified output of the diodes 11r is applied to the PIN diode 11 via transistor switch lit and bias resistor R. A base emitter control current is supplied to transistor lit via resistors 12b and 12e. If a strong radiation field falls on the antenna, a micro-wave voltage is produced across the diode 11r and the current which is aligned in the same way charges the capacitor C. This produces a control current for the diode 11 via basic resistor R and transistor lit. Transistor lit amplifies the driving current, which is therefore small compared to the current absorbed by the diode lis when it is in a conductive state.
    Another way of applying the direct voltage and the control signals is via metal tracks, for example track 29 as indicated in Fig. 6. These metal tracks can be placed at various locations around the antenna metal 3, 3 '. Since they are capacitively coupled to the antenna metal, they will always divert part of the antenna current, with the result that the desired re-radiated power is disturbed or dissipated to a certain extent. However, the micro-wave impedance of the tracks 29 can be increased, at least over a narrow bandwidth, by including, for example, meanders 31 and capacitors 33 as resonance stops. An increase in impedance reduces the microwave currents in the tracks and therefore has the effect of reducing the efficiency loss.
    An FM phase modulator with a single crossed dipole reflector 3 is shown in FIG. This modulator consists of a dielectric lens 41 on the rear surface of which the crossed dipole 3 is mounted. The lens 41 comprises within its construction a selective polarization mirror 43. A transmitting dipole 45 borders on the side of the lens 41 and illuminates the element 1 in cooperation with the mirror 43. The crossed dipole 3 in particular has a reactive load consisting of a number of impedances to be selected with the aid of switches, in combination with a cooperating logic function circuit to enable a three-bit phase shift selection. The crossed dipole 3 is joined with its composite dipoles at an angle of 45 ° to the plane of polarization of the incident radiation directed from the transmitting dipole 45. The load impedances are selected such that the re-radiated field is perpendicularly polarized. The radiation directed from the phase control element therefore passes through the mirror 31 without any noticeable reflection occurring. Phase shifts of 0, ft / 4, 7 p / 2, 3 ph * / 4, TC, 51C / U, 3K / 3, 7'K / 4 can be selected and inserted under the three-bit logic control to achieve a to provide step-by-step discrete phase modulation. These phase shifts can be provided at least approximately by three switchable diode capacitor series circuits (11s / 11c in FIG. 1). Since the phase is not a linear function of the capacitance, the aforementioned phase shift intervals It * / 4 will not be accurately reproduced. If precise phase shift intervals of TC / 4 are required, seven diode-capacitor combinations would be required, at least one of which is in a conductive state.
    Systems can be constructed with a plurality of single or crossed dipoles and using a common substrate. The phase introduced at each dipole position can then be controlled for a variety of applications, for example for beam direction control. An example of such an application is shown in FIG. Here, a system 47 consisting of four single or crossed dipoles 48 is arranged on the rear surface of a dielectric lens 49. Radiation is directed at the system from a dipole emitter 45. Microwave power is again radiated through the system and focused according to a beam through lens 49. The position of the virtual image I of the transmission dipole 45 can be varied and thus the beam direction can be controlled by appropriate phase insertion in each of the dipoles 48.
    Another form of a phase control element 1 with a crossed dipole is shown in FIG. In this construction form, the load impedance over one of the two dipoles 3, 3 'can be varied by the radiation transducer level, rather than by applying a bias voltage from an external circuit, as discussed above. The polarization of the radiation reflected by this phase control element 1 differs for high and low power radiation levels. The impedance network 7 connected between the two constituent legs 3a, 3b of one of the dipoles 3 comprises an anti-parallel pair of diodes lis and 13s, that is to say that these diodes are connected in parallel over the gap between the two legs 3a and 3b and so are connected that the polarity of one of the diodes is the opposite of that of the other diode 13s. The diodes lis and 13s can be of the same type, for example both can be Schottky barrier diodes.
    The diodes lis and 13s can also be of different types, for example one diode lis can be a Schottky barrier diode and the other diode 13s a PIN diode. If the power level of the incident radiation is low, both diodes lis and 13s are non-conductive and the network 7 represents a high impedance load for the dipole 3. However, if the power level of the incident radiation is high, both diodes lis and 13s are conductive, so that the load impedance of the network 7 drops to a low value compared to the dipole impedance. The phase of the radiation reflected by this dipole 3 thus differs approximately 11 for low and high power levels of the radiation. The second dipole 3's has an open circuit load and is perpendicular to the first dipole 3. At a low power level, the two dipoles 3, 3 'are loaded accordingly. The radiation plane that is polarized at TC / 4 for the two dipoles 3, 3 "is reflected with unchanged polarization. At high power levels, however, the dipole loads will differ and, in the ideal situation, the radiation reflected by a dipole 3 will be TT be out of phase with that reflected by the other dipole 3 '. In the practical situation, however, the phase difference will be approximately equal to 1C. The plane of the incident radiation polarized in parallel with the axes X or Y shown energizes both dipoles 3, 3 'equally because the dipoles 3, 3' are aligned according to / 4 or -IC / 4 with respect to the axes X, Y . The reflected radiation is flatly polarized, but parallel to the perpendicular axes Y and X, respectively, due to the phase shift.
    A variation of this latter construction form is shown in FIG. Here a load 7 'with a low impedance, for example a short circuit, is connected between the legs 3'a, 3'b of the second dipole 3'. In this case, the reflected radiation is polarized in a direction perpendicular to the incident radiation at low power levels if the diode impedance is high, and in parallel with the incident radiation if the diode impedance is low. As is customary in microwave theory, open chains and short circuits are treated and considered as extreme cases of reactive loads.
    A system of such crossed dipoles according to FIGS. 9 or 10 can be used in a radar installation for coupling a transmitter source and one or more receivers to a common aperture. An example of a duplex radar is shown in FIG. This radar comprises a body of dielectric material 5 with a front face 5a shaped as a dielectric lens. This radar also comprises a system 1 of crossed dipoles as shown in Fig. 9 and receiver Rx and a transmitter Tx placed next to the respective surfaces 5b, 5c and 5d of the dielectric body 5. The surfaces 5c and 5d are mutually perpendicular and both hells at an angle of 1 £ / 4 with respect to the surface 5b. The body 5 comprises a tilted selective polarization mirror 43. The mirror 43 is formed by vapor-deposited parallel metal strips on an exposed surface of the body 5 (not shown), the center lines between the strips being less than tlf / 4 and the strip width being less then the gap between the strips. It is necessary that the body is originally produced in two separate parts (not shown) in order to be able to mount the mirror before joining. Low power level radiation falling on the surface 5a is focused to the receiver Rx. However, this radiation is first converged and reflected at the set of control elements 1 and then reflected a second time at the selective polarization mirror 43. The polarization of the signal radiation remains unchanged. The transmitter source Tx is oriented such that radiation is sent into the dielectric body 5 with a polarization such that it can pass through the mirror 43. (The output radiation of the transmitter and the reflected incident radiation have a mutually perpendicular polarization at mirror 43). The transmitter output radiation is of a high power level. When the transmitter output radiation is reflected by the system 1 of crossed dipoles shown in FIG. 9, the polarization is rotated by 1/2. The output radiation leaving the surface 5a is therefore polarized in parallel with the incoming signal radiation.
    A duplex radar can be constructed differently using phase control elements shown in FIG. In this case, either Rx and Tx are swapped in position compared to the embodiment of FIG. 11, or the selective polarization mirror 43 is oriented such that its metal strips are perpendicular to those of FIG. 11. The polarization of the transmitter output radiation remains then unchanged, while the polarization of the incident signal radiation changes upon reflection by the system. As in the previous example, the outgoing radiation is polarized in parallel with the incoming radiation.
    Fig. 12 shows a further phase control element 50 according to the invention. The element 50 has two dipole legs 51a and 51b connected to the arms 52a and 52b respectively of a short transmission line 52. A variable diode 53 connects the dipole legs 51a and 51b across the width of the arms 52a and 52b, while a capacitor 54 closes the transmission line 52 . A second transmission line 55 with arms 55a and 55b and provided with resistors 56a and 56b is connected to the short transmission line 52 and provides the bias voltage to be applied to varactor 53. The resistors 56a and 56b prevent microwave power loss in line 55.
    The device according to FIG. 12 works as follows. The susceptance of the variator diode 53 at the microwave frequency depends on the bias voltage across this diode and also on the strength of the microwave voltage. The phase of the radiation emitted again by the element 50 is therefore controlled by the bias voltage across the variable factor 53 for the reason stated above. The phase will depend to a certain extent on the strength of the incident microwave power because the varactor tolerance varies with the microwave voltage. The phase will be fully determined by the bias under two conditions: either (a) the microwave voltage is very small as when the phase control element 50 is used in a microwave receiver, or (b) the microwave power level is a fixed quantity which is the case as the phase control element 50 is used in a transmitter. For practical applications, the phase is therefore driven by the bias across the varaktor.
    Reference is now made to Fig. 13 where a phase control element 60 with a crossed dipole is shown. This is the equivalent of a pair of crossed elements 50 and contains dipoles 61 and 61 'with the legs 61a, 61b, 61'a and 6l'b. These legs have respective slots 62a, 62b, 62'a and 62'b for providing transmission lines, the latter being determined by capacitors formed by superimposed plates 63a, 63b, 63'a and 63'b. Four variator diodes 64a, 64b, 64'a and 64'b are connected between the dipole legs as shown, thereby bridging the slots 62a, 62b, 62'a and 62'b, respectively. The polarities of the variable diodes correspond to a bridge rectifier circuit. The diode voltage voltage terminals 65a, 65b, 65'a and 65'b are used and include resistors 66a, 66b, 66'a and 66'b for reducing microwave power loss, respectively.
    The phase control element 60 with crossed dipole operates as follows. The load applied to the dipole 61 includes the sealed transmission lines formed by slotted dipole legs 61'a and 61'b, together with varicators 64'a and 64'b. The variable factors 64'a and 64'b are preferably equal in the sense that they have the same dependence on the capacitance on the voltage. In addition, it is preferable to keep the bias voltages across the variable tower 64'a and 64'b. As a result, the microwave currents through these two variables will be the same if the microwave voltages are the same. The radiation incident on and polarized in parallel with the dipole 61 causes currents therein and these will be equally distributed between the variance tower 64'a and 64'b. No microwave voltage will be generated across varario tower 64a and 64b. For the reasons described above in the circuit according to FIG. 12, therefore, the bias voltage across the variance factors 64'a and 64'b controls the phase of the radiation retransmitted by dipole 61 with respect to that of the incident radiation. The variable factors 64a and 64b are preferably also the same and their bias voltages are preferably also the same. The bias voltage over these variors therefore controls the re-irradiation phase through dipole 61 ', relative to that of the incident radiation which is polarized in parallel with dipole 61'. If the bias voltage applied to bias connections 65a, 65b, 65'a and 65'b are equal to Vj + V2, 0, V2, respectively, and the d.c. b equal to V ^. The supply of the bias voltages to these bias connections thus provides independent control of the phase of the retransmitted radiation for the two polarizations.
    Fig. 14 shows a reflecting device 70 for controlling the direction of the emitted radiation. The device 70 comprises multi-element arrays 71 of phase control elements 72a to 72d of four either single or (preferably) crossed diodes mounted on a flat back surface 73 of a plankonvex first dielectric lens 74. The number of elements 72 is not critical. The lens 74 shares a spherical interface 75 with a concave-convex second dielectric lens 76 with an outer surface 77. This device forms a composite lens. If the dielectric constants of the first and the second lens are 6 and 2, respectively, 1 is greater than 6.2 and both are high compared to those of the free space, as will be described. A transmitter 78 is mounted on a third surface 79 of the first lens 74 and is designed such that the system 71 re-radiates to a selective polarization mirror 80 after reflection. The dipoles 72 change the radiation polarization to that emitted by the mirror 80. The radiation is refracted again at the spherical surface 75 between the lenses 74 and 76. The curvature of the interface 75 is such that each of the dipoles 72a to 72d the radiation incident thereon reflects through a respective area 81a to 81d of the second lens which forms the outer surface 77. The regions 81a to 81d are embodied such that they merge into one another substantially as shown. The radiation pathways 82b and 82c are respectively indicated as dotted lines and broken lines for the internal dipoles 72b and 72c. It should be noted that the radiation emanating from the surface 77 of the outer lens is inverted with respect to the dipole position in the system 71.
    The radiation reflected by the system 71 produces a wavefront in the free space (not shown) that leaves the surface 77 of the outer lens, the direction of the wavefront being determined by the relative phases of the contributing radiation that reaches the surface areas 81a to 81d of the outer lens.
    Each contribution will have a phase consisting of a fixed component determined by that of the output of the transmitter 78 and a variable component determined by the operational state (e.g., the bias bias) of the associated dipole 72. Accordingly, beam formation can of the radiation from the surface 77 of the outer lens are carried out by appropriate selection of the dipole loads, for example by switching on suitable cadencers or adjusting suitable variable factor bias as described with reference to FIGS. 1 and 12, respectively.
    This bundle forming technique requires that P 2 (second lens 76) be high compared to that of the free space because two states are required that control the dimensions of the regions 81a to 81d. First, the separation between the centers of these areas must be less than q / 2, where λ q is the radiation wavelength in the free space. Secondly, the distance should not be less than the optical resolution provided by the first and second lenses 74 and 76. This resolution is kAj / 2 sinGj, where k is a number close to 1.2, the wavelength is in the second lens 76, that is
    and is half the angle of the conversion radiation cone illuminating a surface area 98 of an outer lens.
    To meet the above conditions, the refractive index n2 of the dielectric material forming the second lens 76 must be greater than n determined by: n * ^ Ο ^ λ-1 = k / sinGj ^ may be, for example, about 25 °, in which case n = 2.8 and n ^ 8 *> n2 must therefore be greater than 2.8 and £ -2 = n ^ must be greater than 8. In addition, β ^ must be greater than 12 as mentioned earlier. It is not difficult to meet these criteria in practice at microwave frequencies. For example, ceramic aluminum oxide has a dielectric constant (€ 2 ^ 10 and zirconium titanate stannate (ZTS) a dielectric constant (Éj) of ^ - * 36.
    In order to improve the adaptation of the phase control system 71 to the combination of lenses 74 and 76, each of the dipoles 72a to 72d can be provided with a relatively small converging lens. Each small lens can easily be deployed in the rear surface 73 of the first lens 74. The small lenses will be concave or convex depending on their lens materials having dielectric constants that are smaller or larger than ε ^.
    The small or individual lenses for the phase control change the polar diagram of their respective dipoles. The composite polar diagram of the system 71 can accordingly be fine-tuned to a desired configuration by appropriate variation of the individual focusing properties of the small lenses. The incorporation of these lenses provides an additional degree of freedom for optimizing the beam configuration of the phase control system. The optical design is known per se in optics and will not be described in further detail.
    权利要求:
    Claims (21)
    [1]
    Phase-controlled reflector element for microwave radiation, which element is provided with a dipole (3), characterized in that the element is also provided with: (1) a substantially lossless dielectric member (5) placed next to the dipole (3) and such that a strong radiation is coupled to it, and (2) a variable reactance (7) performed as a substantially loss-free load for the dipole (3), whereby radiation incident on the dipole (3) is re-irradiated with a phase variable according to the sign and the size of the loss response.
    [2]
    Reflector element according to claim 1, characterized in that the dipole (3) and the variable reactance (7) are of planar construction.
    [3]
    Reflector element according to claim 1 or 2, characterized in that the variable reactance (7) has a strength that can be controlled by a direct current signal applied thereto.
    [4]
    Reflector element according to claim 3, characterized in that the variable reactance comprises at least one variable diode (53) with bias connections (55a, 55b) for capacitance variation.
    [5]
    Reflector element according to claim 3, characterized in that the variable reactance (7) comprises at least one switchable reactance (11c, 13c).
    [6]
    Reflector element according to claim 4 or 5, wherein the variable reactance (11c, lis) is capacitive and is connected in parallel with a self-inductance (9). i
    [7]
    A reflector element according to claim 8, characterized in that the self-induction is a slotted second dipole (3 ') arranged transversely to the reflector dipole element (3).
    [8]
    A reflector element according to claim 1, characterized in that the dipole is a first dipole (3) disposed transversely of a second dipole (3 ') which provides a combination for coupling with different radiation polarizations via the dielectric member (5).
    [9]
    Reflector element according to claim 8, characterized in that the variable reactive load of the first dipole (3) comprises an anti-parallel pair of diodes (lis, 13s) that exhibit a variable impedance from high to low with a change in the energy level of the incident radiation from low to high.
    [10]
    Reflector element according to claim 8, characterized in that the second dipole (3 ') has a substantially loss-free load provided with a second variable reactance (7', 11 ', 13').
    [11]
    Reflector element according to claim 10, characterized in that the first and the second dipole (3, 3 ') are each provided with slots for providing an inductive contribution to the other variable reactance, each variable reactance also being a respective one. variable capacitive element (11, 13, 1Γ, 13 ').
    [12]
    The reflector element according to claim 11, wherein the capacitive elements (11, 13, 11 ', 13') can be selected by switches.
    [13]
    The reflector element according to claim 1, characterized in that the dipole (3'a, 3'b) is disposed between a layer of substantially lossless semiconductor material (21) and the dielectric member (5).
    [14]
    Reflector element according to claim 13, characterized in that the layer of semiconductor material (21) has an associated metal layer (22) which is arranged at a distance from the dielectric member (5).
    [15]
    A reflector element according to any one of the preceding claims, characterized in that the dipole (44) is designed as a member of a set (47) of similar dipoles (44).
    [16]
    A reflector element according to claim 15, characterized in that the system (47) is designed for reflecting radiation from a source (45) through a lens (49).
    [17]
    The reflector element according to claim 1, characterized in that the dipole (3) is crossed by a second dipole and is arranged to receive radiation from a source (45) after reflection by a polarization-selective mirror (43), the dipole (3) and the second dipole are configured such that the polarization of the incident radiation changes and reflects it for transmission through the mirror (43).
    [18]
    The reflector element of claim 1, wherein the dipole (3) is crossed by a second dipole (3 ') and is configured to receive radiation either from the free space or from a source (Tx) after transmission through a polarization-selective mirror ( 43), wherein the dipole (3) and the second dipole (3 ') are configured such that the polarization of the incident radiation changes and reflects it either for reflection through a polarization-selective mirror (43) or to a receiver (Rx) or to the free space.
    [19]
    The reflector element according to claim 1, characterized in that (1) the dipole (72) is designed as a member of a system (71) of dipoles (72a to 72d), each with a respective variable reactive load whose strength is due to an applied bias voltage can be controlled, (2) wherein the dielectric member is embodied as a lens (74) provided with a polarization-selective mirror (80) and cooperates with a second lens (76) with a lower dielectric constant, which is large compared to that of the free space, (3) a transmitter (78) is designed such that radiation is directed to the mirror (80) for reflection to the system (71), (4) the system (71), the mirror (80) ) and the lenses (74, 76) are configured such that the radiation reflected by the system (71) is emitted from the mirror (80) and passes through the lenses (74, 76), each dipole (72) reflecting the radiation - lives through a respective outer surface area (81) of the two the lens (76).
    [20]
    A reflector element according to claim 19, characterized in that each dipole in the system is crossed by a respective second dipole.
    [21]
    A reflector element according to claim 1, characterized in that the dipole (61) is crossed by a second similar dipole (61 '), each of the dipoles (61, 61') being slotted and designed to form of an inductive load for the others, and in which the dipoles (61, 61 ') have variable reactive loads consisting of respective variable diodes (64a, 64b, 64'a, 64'b).
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    同族专利:
    公开号 | 公开日
    IT8547728D0|1985-02-26|
    GB2237936B|1991-10-02|
    FR2685550A1|1993-06-25|
    DE3506933C2|1995-04-13|
    IT1227287B|1991-04-04|
    NL194934B|2003-03-03|
    GB2237936A|1991-05-15|
    FR2685550B1|1995-03-03|
    CA1295417C|1992-02-04|
    NL194934C|2003-07-04|
    DE3506933A1|1991-10-31|
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    法律状态:
    2003-02-03| A1C| A request for examination has been filed|
    2005-05-02| V4| Lapsed because of reaching the maximum lifetime of a patent|Effective date: 20050226 |
    优先权:
    申请号 | 申请日 | 专利标题
    GB8405309|1984-02-27|
    GB8405309|1984-02-27|
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