• 专利附录
  • 专利说明
  • 权利要求
  • 类似技术
  • 同族专利
  • 引用文献
  • 法律状态
  • 优先权
    • 专利摘要:
    An electronic system comprising a first (OSC1) and a second (OSC2) oscillator mutually coupled and having the same resonance frequency, each said oscillator comprising an electric resonator (RES1, RES2), an active cell (CA1, CA2) having a resistance of small negative signal connected to said electric resonator, a power supply terminal (AE1, AE2) of said active cell, an output (S1, S2) for an oscillation signal (s_osc1, s_osc2) and a connection terminal (CM1, CM2) to a mass point, characterized in that: - the power supply terminal (AE2) of the second oscillator and the terminal (CM1) of connection to a ground point of the first oscillator are connected to a same point (MD ), called dynamic mass; and the system also comprises a differential amplifier (CMFB) forming, with the active cell of one of said oscillators, a feedback loop adapted to maintain the potential of said dynamic mass point at a constant level, a function of said voltage reference.
    公开号:FR3067890A1
    申请号:FR1755426
    申请日:2017-06-15
    公开日:2018-12-21
    发明作者:Baudouin Martineau
    申请人:Commissariat a lEnergie Atomique CEA；Commissariat a lEnergie Atomique et aux Energies Alternatives CEA；
    IPC主号:
    专利说明:

    The invention relates to an architecture allowing voltage sharing between two electronic oscillators. It relates to the field of analog electronics, and in particular to radio frequency (RF) electronics, and applies in particular to the production of voltage-controlled oscillators (VCO, for “Voltage Controlled Oscillator”) for RF transmitters or receivers.
    The downscaling of silicon technologies carried out in recent years has led to a sharp reduction in the supply voltages of the transistors, as well as the maximum voltage that can be applied across their terminals. This results in a strong limitation of the maximum admissible amplitude of the signals that can be generated or processed by circuits using these transistors. For example, CMOS transistors in 65 nm technology are designed to support a drainsource voltage of 1.32 V. These transistors are generally supplied at 1.2 V; in the case of a crossed differential pair type oscillator this would lead to a maximum drain-source voltage reaching about 2.2 V at the positive peak of the oscillation. In order to ensure reliable operation, it is therefore necessary to limit the supply voltage to approximately 700 mV, and therefore the peak amplitude of the oscillation signal to only ± 500 mV. For example, in the article by J. L. Gonzalez et al. "A 56-GHz LC-Tank VCO With 7% Tuning Range in 65-nm Bulk CMOS for Wireless HDMI", RFIC 2009 - IEEE Radio Frequency Integrated Circuits Symposium, 2009, describes a crossed differential pair CMOS oscillator in which a PMOS transistor is used to supply a bias current at a voltage level lower than that of the circuit supply.
    This results in a significant reduction in the signal to noise ratio (SNR, from the English "Signal to Noise Ratio"), due in particular to phase noise. This affects the spectral purity of the oscillation signal, which is undesirable especially in telecommunications applications.
    A first approach to address this problem is microelectronic technology. The founders have indeed introduced new families of transistors: double gate oxide transistor (GO2) and extended drain transistor (LDMOS) for CMOS technologies, high voltage transistor in bipolar technology. In doing so, they were able to maintain voltages in the range of 1.5V to 5V at the cost of increased manufacturing costs. For example, STMicroelectronics' UTBB FDSOI 28nm technology has both single oxide (maximum voltage 1V) and double oxide (maximum voltage 1.5-1.8V) transistors. However, these voltages (1.5-1.8V) remain insufficient to reach the SNR levels required in current RF systems.
    A second approach, alternative or complementary to the first, consists of designing assemblies allowing a supply voltage that is too high to be distributed between several transistors to be directly applied to the terminals of a single device. For example, the article by J. Dang et al "A fully integrated 5.5 GHz cross-coupled VCO with high output power using 0.25pm CMOS technology" 2014 21 st IEEE International Conférence on Electronics, Circuits and Systems (ICECS), Marseille, 2014, pp. 255-258 describes an oscillator comprising a first differential pair of crossed PMOS transistors and a second differential pair of crossed NMOS transistors. The sources of the PMOS transistors of the first pair are connected to a power supply, and their drains to those of the transistors of the second pair and to the terminals of a parallel LC resonant circuit. The sources of the transistors of the second pair are connected to ground. In this way, the supply voltage is shared between the two differential pairs, which makes it possible to obtain an oscillatory signal of higher amplitude than if only one pair was used. The disadvantage of this approach lies in the need to use PMOS transistors, whose performance - particularly in terms of noise - is less good than that of NMOS transistors.
    Another possibility is to use transistors in "cascode" configuration. This solution, however, imposes frequency limitations and degrades the performance of the circuit.
    The invention aims to overcome the aforementioned drawbacks of the prior art.
    An object of the invention, enabling this object to be achieved, is an electronic system comprising a first and a second oscillator mutually coupled and having the same resonant frequency, each said oscillator comprising an electric resonator, an active cell having a resistance of small negative signal connected to said electrical resonator, an electrical supply terminal of said active cell, an output for an oscillation signal and a connection terminal to a ground point, characterized in that:
    the power supply terminal of the second oscillator and the connection terminal to a ground point of the first oscillator are connected to the same point, called dynamic ground; and the system also includes a differential amplifier having a first input which can be connected to a reference voltage source, a second input connected to said dynamic ground point and an output connected to a terminal for controlling the bias current of one of the first and second oscillators so as to form a feedback loop adapted to maintain the potential of said dynamic ground point at a constant level, a function of said reference voltage.
    According to particular embodiments of the invention:
    The system may also include a signal combiner circuit configured to add the voltages of the oscillation signals taken from the outputs of the first and second amplifiers.
    More particularly, this signal combining circuit can comprise a transformer comprising a first and a second primary winding, respectively connected to the outputs of the first and the second oscillator, and two secondary windings connected in series and coupled respectively to the first and to the second primary winding.
    Said primary windings can in particular be inductances of the electrical resonators of the oscillators.
    The power supply terminals of said first and second oscillators can be connected to respective midpoints of said primary windings.
    The system may also include said reference voltage source, configured to generate said reference voltage of a value equal to half that of said supply voltage.
    The electrical resonators of said first and second oscillators can be parallel LC circuits.
    The active cells of said first and second oscillator may each comprise at least one transistor having a gate or base, the output of said differential amplifier being connected to the base of the transistor of the active cell of the first oscillator.
    The active cells of said first and second oscillators may each comprise a differential pair of crossed transistors. In this case, the output of said differential amplifier can be electrically connected, in direct current, to gates or bases of the crossed transistors of the differential pair of the active cell of said first oscillator, and the gate or base of each of said crossed transistors of the differential pair of the active cell of said first oscillator can be connected to the drain or collector of the other transistor of the pair via a decoupling capacitor.
    Said oscillators can be voltage-controlled oscillators, configured to be controlled together so as to have the same oscillation frequency.
    The system may also include a source of direct supply voltage connected to the power supply terminal of said first oscillator.
    Said oscillators can be radio frequency oscillators.
    Said oscillators can be co-integrated monolithically.
    Other characteristics, details and advantages of the invention will emerge on reading the description made with reference to the accompanying drawings given by way of example and which represent, respectively:
    FIG. 1A, a diagram of an electronic system according to an embodiment of the invention;
    Figures 1B and 1C, the electrical diagrams of the two oscillators of the system of Figure 1A;
    FIG. 2A, a graph of the shape of the oscillation signal generated by the oscillator of FIG. 1C taken in isolation;
    Figure 2B, a graph of the spectral density of the phase noise of the signal of Figure 2A;
    Figure 2C, the signal spectrum of Figure 2A;
    FIG. 3A, a graph of the shape of the oscillation signals generated by the two oscillators of the electronic system of FIG. 1A;
    FIG. 3B, a graph of the shape of the oscillation signal at the output of the electronic system of FIG. 1A;
    Figure 3C, a graph of the spectral density of the phase noise of the signal of Figure 3B; and
    Figure 3D, the signal spectrum of Figure 3B.
    The invention is based on the principle of sharing a relatively high supply voltage between two oscillators operating simultaneously, the signals of which can be constructively combined in order to improve their signal-to-noise ratio. This sharing is ensured and stabilized by a feedback loop.
    FIG. 1A shows a diagram of an electronic system according to an embodiment of the invention. This system comprises two electronic oscillators OSC1, OSC2 having the same resonant frequency, the electrical diagrams of which are illustrated in FIGS. 1B and 1C, respectively.
    Each of the two oscillators OSC1, OSC2 comprises an electrical resonator (RES1 for OSC1, RES2 for OSC2) and an active cell (CA1 for OSC1, CA2 for OSC2) with a small negative signal resistance, making it possible to compensate for the losses of the resonator. In the embodiment considered here, the resonators RES1, RES2 are parallel LC circuits: the resonator RES1 comprises an inductance L1 mounted in parallel to a capacitive element C1 and the resonator RES2 comprises an inductance L2 mounted in parallel to a capacitive element C2 .
    In FIG. 1A, the inductors L1, L2 are represented outside the blocks symbolizing the oscillators OSC1, OSC2 to allow a better understanding of the operation of the electronic system.
    In many applications, it is desired that the oscillators OSC1, OSC2 have a variable resonant frequency, and in particular be VCOs. In this case, the capacitive elements C1, C2 are typically PN junctions inversely polarized by a voltage S_VCO which determines their capacity. It is important that the capacitive elements C1 and C2 are controlled jointly in order to maintain the equality of the resonant frequencies of the two resonators.
    The active cells CA1, CA2 are of the type with differential pair of crossed transistors. Thus the active cell CA1 of the first oscillator OSC1 comprises two identical MOSFET transistors, T11 and T12, whose sources S_T11, S_T12 are connected together and the drains D_T11, D_T12 are connected to opposite terminals of the inductance L1 and the element capacitive C1. The gate G_T 11 of the transistor T11 is connected (via a decoupling capacitor CD1 whose function will be explained below) to the drain of the transistor T12, and the gate G_T12 of the transistor T12 is connected (via the 'a decoupling capacity CD2) to the drain of transistor T11. Similarly, the active cell CA2 of the second oscillator OSC2 comprises two identical MOSFET transistors, T21 and T22, whose sources S_T21, S_T22 are connected together and the drains
    D_T21, D_T22 are connected to opposite terminals of the inductor L2 and the capacitive element C2. The gate G_T21 of the transistor T21 is connected (directly, without decoupling capacity) to the drain of the transistor T22, and the gate G_T22 of the transistor T22 is connected (also without the decoupling capacity) to the drain of the transistor T21.
    A direct supply current l dd is injected into the drains of the transistors of each active cell. In the embodiment described here, this current is supplied via a terminal AE1, AE2 connected to the midpoint of the inductance L1, L2 of the oscillator.
    As can be seen in FIG. 1A, a source SVA of direct supply voltage is connected to the supply terminal AE1 of the oscillator OSC1 to maintain it at a potential V dd . The current l dd , coming from the source SVA, enters the drains D_T 11, D_T 12 of the transistors T11, T12 of this oscillator and exits through their sources S_T11, S_T12. The latter are connected to a point MD which is called a "dynamic mass" because its potential does not vary during normal operation of the oscillators. The power supply terminal AE2 of the second oscillator OSC2 is also connected to this point MD. In this way, the current l dd also enters the drains D_T21, D_T22 of the transistors T21, T22 of this oscillator and exits through their sources S_T21, S_T22. These are connected to a ground in the electronic system.
    It is easily understood that the average drain-source voltage of the transistors T11, T12 of the oscillator OSC1 is equal to V dd -V MD , where V MD is the voltage of the dynamic ground point MD, and that of the transistors T21, T22 of the oscillator OSC2 is worth Vmd- Thus, in order to ensure a constant distribution of the supply voltage between the two oscillators, it is necessary to stabilize the voltage V MD .
    According to the invention, this stabilization is obtained by means of a common mode feedback amplifier (CMFB
    Common Mode Feed-Back Amplifier). This type of amplifier is well known in the literature and can, for example, be produced from an operational amplifier. The amplifier CMFB receives a direct voltage, called the reference voltage (Vref) on one of its inputs ED1 (non-inverting input "+") and the voltage V MD on the other input ED2 (inverting input "-"), and provides at its output a signal s_cmbf proportional to the difference between these two voltages. This signal, the frequency of which is much lower than the resonance frequency fo of the oscillators, is applied to the gates G_T 11, G_T 12 of the transistors T11, T12 of the active cell CA1 of the first oscillator OSC1 through resistors RD1, RD2. The decoupling capacities CD1, CD2 are necessary because the drains of these transistors are maintained at an average voltage V dd , different from s_cmbf. The values of these decoupling capacitors, and those of the resistors RD1, RD2, are chosen so that the decoupling capacitors behave like open circuits with respect to the low frequency signal s_cmbf, and like short circuits with the resonant frequency fo, allowing the establishment of an oscillation.
    This arrangement forms a feedback loop guaranteeing that V M d = Vref- Indeed, if V MD decreases and becomes less than V RE f, the voltage level of the signal s_cmbf - and therefore the gate voltage of the transistors T11 and T12, increases. In addition, as Vmd voltage is applied to the gates of transistors T21, T22, the current I d browsing both oscillators also tends to decrease. These two effects cooperate to cause an increase in the source voltage of T11 and T12. But this source voltage is nothing other than Vmd, which thus approaches Vref · Conversely, if V M d increases and exceeds V RE f, the feedback loop intervenes to reduce it.
    The operation of the system can be described in summary as follows. The two oscillators are connected in series in continuous mode: they are therefore crossed by the same bias current l dd . The oscillator OSC2 is self-polarized and imposes a relationship between the value of l dd and that of the potential V M d- The signal s_cmbf is applied to a control terminal of the bias current flowing through OSC1 (a reference point P in the figure 1A - connected to the gate of transistors T11 and T12); this signal determines the value of l dd . The feedback loop fixes the value of s_cmbf, and therefore of l dd , so that the potential Vmd takes the desired value, equal to (or, more generally, function of) V REF .
    In the example of FIG. 1A, VR EF = V dd / 2, which implies Vm = V dd / 2. This means that the supply voltage is distributed equally between the two oscillators. This choice is optimal if the oscillators (or at least their transistors) are identical, but it is quite possible to distribute the voltage unevenly by choosing V REF ^ V dd / 2, in particular if the transistors of the two oscillators have channels of different widths.
    In the example of FIG. 1A, the reference voltage V REF is generated by a voltage source SVR which may simply be a voltage splitter connected between an output of the power source SVA and the ground, but of other embodiments are possible.
    The oscillators OSC1, OSC2 generate oscillation signals s_osc1, s_osc2 of the same frequency, the amplitudes of which depend on the average drain-source voltages of the transistors of said oscillators. Thus the amplitude of the signal s_osc1 generated by the oscillator OSC1 depends on V dd -V REF , while that of the signal s_osc2 generated by the oscillator OSC2 depends only on V REF . The amplitudes are equal if the oscillators are identical and V REF = V dd / 2. These two signals can be taken from output gates S1, S2 whose terminals are connected to the drains of the transistors of the respective oscillators or, in an equivalent manner, across the terminals of the inductors L1, L2 and the capacitive elements C1, C2. They can then be combined, that is to say summed, to obtain an output signal s_out of higher amplitude, therefore less subject to phase noise. In the embodiment of FIG. 1A, this is obtained by means of a transformer TS having two secondary windings ES1, ES2 connected in series and inductively coupled to the inductors L1, L2 which play the role of primary windings.
    In order for the combination of the two oscillation signals to lead to the generation of an output signal of greater amplitude, it is necessary that the two oscillators are tuned (therefore have the same oscillation frequency) and synchronized (it that is, they oscillate at least approximately in phase). In practice, this is obtained automatically thanks to the necessarily present mutual couplings. These couplings are due on the one hand to the transformer TS, and on the other hand to the physical proximity of the oscillators, especially when they are co-integrated in a monolithic manner.
    The article by S. A. R. Ahmadi-Mehr, M. Tohidian and R. B. Staszewski, Analysis and Design of a Multi-Core Oscillator for Ultra-Low Phase Noise, IEEE Transactions on Circuits and Systems I: Regular Papers, vol. 63, no. 4, pp. 529-539, April 2016, describes a system exploiting the coupling between two electronic oscillators.
    FIG. 2A shows the shape of an oscillation signal of an oscillator of the type of FIG. 1C having a supply voltage of 1.2 V. The signal is sinusoidal, with a frequency f 0 1 = 17, 12 GHz. FIG. 2B shows that this signal is affected by phase noise, the power spectral density of which relative to the signal power at 17.12 MHz (and measured in dBc / Hz) is illustrated in FIG. 2B. We can see that, at 1 MHz, the noise level is around -99.4 dBc / Hz. This is sufficient to affect the spectral purity of the signal, as can be seen in Figure 2C.
    FIG. 3A shows the two oscillation signals, s_osc1 and s_osc2 generated by an electronic system of the type of FIGS. 1A - 1C; these signals have the same frequency and a constant phase relationship, even if they are not exactly in phase with each other. Due to the presence of the TS transformer and the mutual coupling of the oscillators, the frequency of these signals is slightly modified compared to that of an oscillator alone, and is equal to fo = 18.85 MHz. FIG. 3B shows the shape of the signal s_out obtained by combining the two individual signals s_osc1 and s_osc2: it can be observed that the peak-to-peak amplitude of s_out is almost 4V. Figure 3B shows that the phase noise has a power spectral density of around -104.4 dBc / Hz, an improvement of 5 bBc / Hz in the phase noise (theoretically, we would expect a gain of 6 dBc / Hz, but the transformer has losses and the synchronization of the oscillators is imperfect). FIG. 3D makes it possible to verify that the spectral purity of the signal is significantly improved.
    The invention has been described with reference to a particular embodiment, but many variants are possible. For example :
    The feedback signal s_cmbt could be applied to the second OSC2 oscillator. In this case, the voltage V MD should be applied to the non-inverting input of the differential amplifier and the reference voltage to its inverting input.
    The oscillators OSC1, OSC2 need not be VCOs.
    Oscillators do not necessarily have to operate in the radio frequency range (1 MHz or more), or even microwave frequencies (1 GHz or more): the invention is also suitable for oscillators operating at a lower frequency.
    The active cells of the oscillators need not necessarily be of the cross differential pair type; other circuit topologies may provide small negative signal resistance. This is the case, in particular, of the classic "Colpitts" and "Hartley" assemblies with a single transistor. The use of a differential structure is nevertheless preferred.
    Electric resonators do not have to be of the parallel LC type. If the resonant frequency can be fixed, it can even be crystal resonators.
    The oscillator transistors need not be of the MOSFET type. It can in particular be bipolar transistors. It is interesting to note that the bipolar transistors are subject to a risk of thermal runaway. To avoid this, it is common to connect a so-called "ballast" resistor to the emitters of the bipolar transistors in the common emitter configuration; this however has the disadvantage of degrading the energy efficiency of the circuit. In the case of the invention, the feedback loop which stabilizes the value of V MD is sufficient to prevent any thermal runaway, even in the absence of ballast resistors. The energy efficiency of oscillators in bipolar technology is improved.
    It is advantageous to inject the bias current l d d 5 in correspondence with the midpoints of the inductances of the oscillators. This, combined with the use of differential type active cells, prevents oscillation signals from traveling through the feedback loop, disrupting the functioning of the system. Alternatively, it is possible to use capacitors connected to ground to filter the oscillation signals in the feedback loop.
    The signals generated by the two oscillators can be combined by means other than a transformer, for example a Wilkinson divider used as a combiner.
    权利要求:
    Claims (14)
    [1" id="c-fr-0001]
    1. Electronic system comprising a first (OSC1) and a second (OSC2) oscillator mutually coupled and having the same resonant frequency, each said oscillator comprising an electric resonator (RES1, RES2), an active cell (CA1, CA2) having a small negative signal resistance connected to said electrical resonator, a power supply terminal (AE1, AE2) of said active cell, an output (S1, S2) for an oscillation signal (s_osc1, s_osc2) and a connection terminal ( CM1, CM2) at a mass point, characterized in that:
    the power supply terminal (AE2) of the second oscillator and the terminal (CM1) of connection to a ground point of the first oscillator are connected to the same point (MD), called dynamic mass; and the system also includes a differential amplifier (CMFB) having a first input (ED1) which can be connected to a reference voltage source (SVR), a second input (ED2) connected to said dynamic ground point (MD) and an output (SD) connected to a terminal (P) for controlling the bias current of one of the first and second oscillators so as to form a feedback loop adapted to maintain the potential of said dynamic ground point at a level constant, function of said reference voltage.
    [2" id="c-fr-0002]
    2. The electronic system according to claim 1 also comprising a signal combiner circuit (TS) configured to add the voltages of the oscillation signals taken from the outputs of the first and second amplifier.
    [3" id="c-fr-0003]
    3. The electronic system as claimed in claim 2, in which said signal combiner circuit comprises a transformer (TS) comprising a first (L1) and a second (L2) primary winding, respectively connected to the outputs of the first and of the second oscillator, and two windings. secondary (ES1, ES2) connected in series and coupled respectively to the first and second primary winding.
    [4" id="c-fr-0004]
    4. The electronic system as claimed in claim 3, in which said primary windings are inductors of the electrical resonators of the oscillators.
    [5" id="c-fr-0005]
    5. Electronic system according to one of claims 3 or 4 wherein the electrical supply terminals (AE1, AE2) of said first and second oscillator are connected to respective midpoints of said primary windings (L1, L2).
    [6" id="c-fr-0006]
    6. Electronic system according to one of the preceding claims, also comprising said reference voltage source (SVR), configured to generate said reference voltage (V REF ) with a value equal to half that of said supply voltage ( Vdd) ·
    [7" id="c-fr-0007]
    7. Electronic system according to one of the preceding claims, in which the electrical resonators of said first and second oscillators are parallel LC circuits.
    [8" id="c-fr-0008]
    8. Electronic system according to one of the preceding claims, in which the active cells of said first and second oscillator each comprise at least one transistor having a gate or base, the output of said differential amplifier being connected to the base of the transistor of the active cell. of the first oscillator.
    [9" id="c-fr-0009]
    9. The electronic system as claimed in claim 1, in which the active cells of said first and second oscillator each comprise a differential pair of crossed transistors (T11, T12; T21, T22).
    [10" id="c-fr-0010]
    10. The electronic system as claimed in claim 9, in which:
    the output of said differential amplifier is electrically connected, in direct current, to gates or bases (G_T11, G_T12) of the crossed transistors of the differential pair of the active cell of said first oscillator; and the gates or base of each of said crossed transistors of the differential pair of the active cell of said first oscillator is connected to the drain (D_T11, D_T12) or collector of the other transistor of the pair via a decoupling capacitor (CD1, CD2).
    [11" id="c-fr-0011]
    11. Electronic system according to one of the preceding claims, in which said oscillators are voltage-controlled oscillators, configured to be controlled together so as to have the same oscillation frequency.
    [12" id="c-fr-0012]
    12. Electronic system according to one of the preceding claims also comprising a source (SVA) of direct supply voltage connected to the power supply terminal of said first oscillator.
    [13" id="c-fr-0013]
    13. Electronic system according to one of the preceding claims, in which said oscillators are radio frequency oscillators.
    [14" id="c-fr-0014]
    14. Electronic system according to one of the preceding claims, in which said oscillators are co-integrated in a monolithic manner.
    类似技术:
    公开号 | 公开日 | 专利标题
    FR3067890B1|2019-08-02|ARCHITECTURE OF VOLTAGE SHARING BETWEEN TWO OSCILLATORS
    EP1916762B1|2018-05-30|Quartz oscillator with amplitude control and an extended temperature range.
    EP0256580A1|1988-02-24|Gyrator simulating an inductance
    EP1424776B1|2006-02-22|Voltage controlled oscillator circuit for a low power electronic device
    EP1265352B1|2011-08-10|Differential oscillator circuit comprising an electromechanical resonator
    EP1909386A1|2008-04-09|Negative-capacitance circuit for high-frequency applications
    CH633670B5|1982-12-31|RECTANGULAR WAVE OSCILLATOR CIRCUIT FOR TIME MEASUREMENT.
    US20070085620A1|2007-04-19|Semiconductor integrated circuit device
    US10454419B2|2019-10-22|Hybrid resonator based voltage controlled oscillator |
    EP2073377A1|2009-06-24|Oscillating quartz circuit with low consumption and active biasing
    US6362698B1|2002-03-26|Low impedance clamping buffer for an LC tank VCO
    US7564316B2|2009-07-21|Variable-frequency oscillator incorporating thin-film bulk acoustic resonators
    US20100102859A1|2010-04-29|Lc voltage controlled oscillator tank with linearized frequency control
    FR3051976A1|2017-12-01|SERIES OF COUPLED SYNCHRONOUS OSCILLATORS
    EP0859458B1|2003-04-09|Circuit for frequency modulation of a quartz oscillator
    FR3059493B1|2019-11-22|REGULATING AN RF AMPLIFIER
    FR2749720A1|1997-12-12|HIGH SPEED RING OSCILLATOR
    FR2549314A1|1985-01-18|ADAPTATION QUADRUPOLE, INDEPENDENT OF OPERATING FREQUENCY, REACTANCE, AND PROGRESSIVE WAVE AMPLIFIER USING SUCH QUADRUPOLE
    FR2904739A1|2008-02-08|COMPENSATION OF AN AMPLIFIER COMPRISING AT LEAST TWO GAIN STAGES
    FR2813481A1|2002-03-01|VHF/UHF broadcasting variable frequency carrier frequency modulator having frequency/phase detector comparing reference/oscillator output and voltage controlled oscillator applied with modulation/modulation index compensation applied.
    EP1187308B1|2004-06-23|Voltage controlled oscillator
    EP0120756B1|1988-06-22|Oscillator comprising high-frequency bandpass amplifier having an adaptable impedance
    EP3293880A1|2018-03-14|Adaptation circuit for low noise amplifier and low noise amplifier including such a circuit
    FR2962612A1|2012-01-13|MULTI-BAND VOLTAGE CONTROL OSCILLATOR WITHOUT SWITCHED CAPACITY.
    FR2586872A1|1987-03-06|Monolithic microwave frequency/radio frequency converter
    同族专利:
    公开号 | 公开日
    EP3416284B1|2021-06-23|
    US20180367097A1|2018-12-20|
    US10374550B2|2019-08-06|
    EP3416284A1|2018-12-19|
    FR3067890B1|2019-08-02|
    引用文献:
    公开号 | 申请日 | 公开日 | 申请人 | 专利标题
    US3305824A|1965-08-09|1967-02-21|Miles N Brooks|Pest repelling device|
    US20080174375A1|2007-01-19|2008-07-24|Takahiro Nakamura|Voltage-controlled oscillator and communication device using the same|
    US20110057732A1|2009-09-10|2011-03-10|Taylor Stewart S|Low phase noise voltage controlled oscillator|
    KR101138487B1|2011-01-25|2012-04-25|인하대학교 산학협력단|Quadrature voltage controlled oscillator using backgate coupling|
    CN102170289A|2011-05-28|2011-08-31|西安电子科技大学|Low-power-consumption orthogonality LC voltage controlled oscillator base on current multiplex|
    US20140132359A1|2012-11-13|2014-05-15|Qualcomm Incorporated|Current reuse voltage controlled oscillator with improved differential output|
    US20150180528A1|2013-03-26|2015-06-25|Stmicroelectronics S.R.I.|Power oscillator apparatus with transformer-based power combining for galvanically-isolated bidirectional data communication and power transfer|
    US6867656B2|2002-06-17|2005-03-15|California Institute Of Technology|Self-dividing oscillators|US10355646B2|2017-12-20|2019-07-16|Globalfoundries Inc.|Power amplifier for millimeter wave devices|
    US20200036284A1|2018-07-26|2020-01-30|Analog Devices Global Unlimited Company|Power isolator exhibiting low electromagnetic interference|
    FR3099667A1|2019-07-29|2021-02-05|Stmicroelectronics S.R.L.|Distributed active transformer voltage controlled oscillator|
    US10747254B1|2019-09-03|2020-08-18|Globalfoundries Inc.|Circuit structure for adjusting PTAT current to compensate for process variations in device transistor|
    法律状态:
    2018-12-21| PLSC| Search report ready|Effective date: 20181221 |
    2019-06-28| PLFP| Fee payment|Year of fee payment: 3 |
    2020-06-30| PLFP| Fee payment|Year of fee payment: 4 |
    2021-06-30| PLFP| Fee payment|Year of fee payment: 5 |
    优先权:
    申请号 | 申请日 | 专利标题
    FR1755426A|FR3067890B1|2017-06-15|2017-06-15|ARCHITECTURE OF VOLTAGE SHARING BETWEEN TWO OSCILLATORS|
    FR1755426|2017-06-15|FR1755426A| FR3067890B1|2017-06-15|2017-06-15|ARCHITECTURE OF VOLTAGE SHARING BETWEEN TWO OSCILLATORS|
    EP18175127.2A| EP3416284B1|2017-06-15|2018-05-30|Architecture for sharing voltage between two oscillators|
    US15/993,227| US10374550B2|2017-06-15|2018-05-30|Architecture for voltage sharing between two oscillators|
    [返回顶部]