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    • 专利摘要:
    The radar comprising at least one network antenna (10) composed of transmission subnetworks and reception subnetworks, a transmission and reception system and processing means: the distribution of subnetworks emission and reception subnetworks is symmetrical both with respect to a vertical axis (5) and with respect to a horizontal axis (6); at least two transmitting subarrays (11, 11 ') symmetrical with respect to said vertical axis (5) are spaced as far apart as possible; - at least two transmitting subarrays (11, 11 ") symmetrical with respect to said horizontal axis (6) are distant from the greatest distance possible - at least two symmetrical receiving sub-networks (12, 12 ') relative to said vertical axis (5) are as far away as possible, - at least two receiving sub-networks (12, 12 ") symmetrical about said horizontal axis (6) are spaced as far apart as possible; a first coding (101, 102) of the wave emitted by said transmission sub-networks being made by frequency shifting of said ramps between the different transmission subnetworks; a second coding (103, 104) of the wave emitted by said transmission sub-networks being realized by phase modulation of the frequency ramp ramp frequency between the different transmission subnetworks.
    公开号:FR3058227A1
    申请号:FR1601552
    申请日:2016-10-27
    公开日:2018-05-04
    发明作者:Pascal Cornic；Stephane Kemkemian；Yves Audic
    申请人:Thales SA；
    IPC主号:
    专利说明:

    (54) fmcw multibeam radar, especially for cars.
    FR 3 058 227 - A1
    The radar comprising at least one network antenna (10) composed of transmission sub-networks and reception sub-networks, a transmission and reception system and processing means:
    - the distribution of the transmission sub-networks and the reception sub-networks is symmetrical both with respect to a vertical axis (5) and with respect to a horizontal axis (6);
    - at least two transmission sub-networks (11, 11 j symmetrical with respect to said vertical axis (5) are distant by the greatest possible distance;
    - at least two transmission sub-networks (11, 11 “) symmetrical with respect to said horizontal axis (6) are distant from the greatest possible distance;
    - at least two receiving sub-networks (12, 12 ') symmetrical with respect to said vertical axis (5) are distant from the greatest possible distance;
    - at least two reception sub-networks (12, 12) symmetrical with respect to said horizontal axis (6) are distant by the greatest possible distance;
    - A first coding (101, 102) of the wave transmitted by said transmission sub-networks being carried out by frequency shift of said ramps between the various transmission sub-networks;
    - A second coding (103, 104) of the wave transmitted by said transmission sub-networks being carried out by phase modulation of frequency ramp to frequency ramp between the different transmission sub-networks.
    Multi-beam FMCW radar, especially for cars
    The present invention relates to a multibeam FMCW radar. It is particularly applicable in the field of radars for motor vehicles. More generally it can be applied to frequency modulated continuous wave radars, says FMCW.
    Speed cameras for cars have been introduced for driving assistance functions, more oriented towards comfort, such as adapting to cruising speed for use on motorways, known as ACC (Adaptive Cruise Control), or "Stop and Go" in urban driving. They use millimeter waves, in particular the band 76-81 GHz.
    Thanks to the evolution of technologies, current applications are also aimed at anti-collision type safety functions, and it is even envisaged in the relatively short term to achieve a fully autonomous vehicle, the perception of the environment being ensured by the association a number of sensors, based on different technologies: radar, video, infrared in particular
    Because of its all-weather capabilities, the radar remains a predominant sensor in this context and its detection and discrimination capabilities must be extended to guarantee the overall reliability of the system. With regard to anti-collision, the radar sensor must in particular be able to distinguish among the fixed objects it detects, those which correspond to elements of road infrastructure, from those which correspond to vehicles immobilized on the track which potentially constitute a risk of collision. In this context, it is particularly fundamental that it does not generate false alarms that can cause braking or an emergency avoidance maneuver, without any real cause, in particular when the vehicle is traveling at high speed. This imposes increased sensitivity and capacity for discrimination, making it possible to understand the situation in front of the vehicle at a long distance, typically greater than 200 m. It may also be necessary to detect roadsides.
    In this context, the detection sensitivity as well as the resolution and the angular localization capacity in the horizontal plane and in the vertical plane must be optimized simultaneously, while the dimensions of the antenna are particularly constrained.
    A technical problem to be solved is in particular to obtain sufficient sensitivity and angular discrimination capabilities, while retaining a simple antenna architecture and limiting the processing volume.
    To date, this problem has not been resolved. From a purely technical point of view, an ideal solution would be for a constrained antenna surface, to have radiating elements or sub-arrays of radiating elements over the whole of the available surface, these radiating elements or networks being supplied individually by active transmitting and receiving modules. This solution would make it possible to optimize the radiation patterns with respect to the angular resolution, while controlling the level of the secondary lobes of these patterns in transmission and in reception simultaneously. Unfortunately, it is not accessible in the context of a millimeter wave application intended in particular for the automobile, for reasons of technology and cost.
    This is why simpler and less efficient solutions are currently being implemented.
    Radars use network antennas possibly comprising several transmission channels and several reception channels, and carry out by digital calculation the formation of several beams in reception. In this case, the transmission is carried out on one or more antenna networks producing a relatively wide beam, typically more than 20 ° in the horizontal plane and 10 ° in the vertical plane, and the reception is carried out simultaneously or sequentially on several sub-networks covering this same angular domain.
    This technique makes it possible to locate the different targets in the horizontal plane, even in the horizontal plane and in the vertical plane, by comparing the signals received on the different beams.
    In some cases, different antenna sub-arrays are switched in time in transmission mode and in reception mode so as to generate a diversity of radiation diagrams for the purpose of measuring the azimuth and the elevation of the targets or increase angular resolution. Such a principle is described for example in Figure 3 or Figure 4 of the publication "Automotive Radar - Status and Trends" by Martin Schneider (Proceeding the German Microwave Conference GeMIC 2005). This switching is done at the expense of the radar waveform efficiency, which is divided by the number of switching. Furthermore, the presence of switches in the hyper frequency chain causes losses which degrade the sensitivity of the radar, and necessitates increasing the frequency band of reception of the radar, which is also likely to degrade the sensitivity by increasing of the noise power received by the receiver.
    In other cases, the transmission is carried out by an antenna transmission sub-network and the reception is carried out simultaneously on several other antenna reception sub-networks. This solution requires distributing the 76 GHz reference oscillator across all the reception channels to perform synchronous demodulation of the received signals, which can only be envisaged on a small number of channels, given the technological difficulties. Such a principle is described for example in FIG. 5 of the publication "Automotive Radar - Status and Trends" by Martin Schneider cited above. The complexity is further increased when the beam formation must be carried out in azimuth and in elevation.
    In some cases, the same subnets are used simultaneously in transmission and reception, as for example described in the article "Millimeter-wave Radar Sensor Based on a Tranceiver Array for Automotive Applications" by Matthias Steinhauer et al, IEEE Transactions on Microwaves Theory and Techniques (vol. 56, Issue 2, Feb. 2008). In such a solution, the transmission reception transmission is very important, and the radar sensitivity is greatly degraded by the leakage of the noise carried by the transmission in the various receivers. Furthermore, if several transmission sub-networks are activated simultaneously, it is necessary to transmit orthogonal waves on these different sub-networks, which is quickly complex if the number of channels is large. Similarly in reception, for the processing to remain simple and robust, it is necessary to process for a given subnetwork only the signal transmitted by this same subnetwork. The overall impact balance is therefore greatly degraded
    An object of the invention is in particular to overcome the aforementioned drawbacks and to allow the obtaining of sufficient sensitivity and angular discrimination capabilities, while retaining a simple antenna architecture and limiting the volume of processing. To this end, the subject of the invention is a continuous wave radar with frequency modulation according to frequency ramps, called FMCW, comprising at least one network antenna composed of transmission sub-networks and reception sub-networks. , a transmission and reception system and processing means, in which:
    the distribution of the transmission sub-networks and the reception sub-networks is symmetrical both with respect to a vertical axis and with respect to a horizontal axis;
    - at least two symmetric emission sub-networks with respect to said vertical axis are distant by the greatest possible distance;
    - at least two symmetric emission sub-networks with respect to said horizontal axis are distant by the greatest possible distance;
    - at least two symmetrical reception sub-networks with respect to said vertical axis are distant by the greatest possible distance;
    - at least two symmetrical reception sub-networks with respect to said horizontal axis are distant by the greatest possible distance;
    - A first coding of the wave transmitted by said transmission sub-networks being carried out by frequency shift of said ramps between the various transmission sub-networks;
    a second coding of the wave emitted by said transmission sub-networks being carried out by phase modulation of frequency ramp to frequency ramp between the various transmission sub-networks.
    In a possible embodiment, in said first coding, a first half of the transmission sub-networks is supplied by a first FMCW waveform and the second half is supplied by the same frequency-shifted waveform, the two halves being symmetrical about said vertical axis.
    In another possible embodiment, in said first coding, a first left half of the transmission subnets is supplied by a first FMCW waveform and the second left half is supplied by the same waveform shifted in frequency, said two halves being symmetrical with respect to the intersection of said vertical axis and said horizontal axis.
    In said second coding, the waves supplying the various transmission sub-networks belonging to the same line, along the horizontal axis, are for example coded by the same phase code.
    In another possible embodiment, in said first coding, a first half of the transmission sub-networks is supplied by a first FMCW waveform and the second half is supplied by the same frequency-shifted waveform, the two halves being symmetrical with respect to said horizontal axis.
    In said second coding, the waves supplying the different transmission sub-networks belonging to the same column, along the vertical axis, are for example coded by the same phase code.
    Said transmission and reception system comprises for example a first waveform generator VCOa generating a first waveform FMCW and a second waveform generator VCOb, synchronous and coherent with the first, generating the other form FMCW wave shifted in frequency. Each of said waveform generators VCOa, VCOb is for example used both for transmission and for synchronous demodulation of the signals on reception. The frequency difference between said waveform generators is for example chosen so that the beat frequencies of the reception signals demodulated by a same waveform generator VCOa occupy disjoint frequency bands according to whether said reception signals come from a transmission from the same VCOa generator or from the other VCOb generator. The signals in reception resulting from the different transmission subnets are for example separated by filtering and by correlation in reception, respectively according to their frequency band and according to their phase modulation code.
    A first two-plane monopulse beam formation comprising a sum channel and two reception channels is for example carried out in transmission by said processing means, on each reception channel associated with a reception sub-network, using the signals coming from any or part of the transmission sub-networks. A second monopulse beam formation in two planes comprising a sum channel and two difference channels is for example carried out on reception by said processing means, by associating all or part of the signals received on all or part of the reception channels. The detection and the angular localization of targets are for example carried out on the basis of the signals resulting from the product of the transmit monopulse beams and the receive monopulse beams.
    The phase modulation code applied is for example a two-phase code having the value 0 or π.
    The phase modulation code applied is for example a Hadamard code.
    In a particular embodiment, said antenna comprises six transmission sub-networks and eight reception sub-networks, said transmission and reception system comprising two integrated circuits each comprising a waveform generator, three transmission channels and four reception channels, the three transmission sub-networks arranged on one side of one of said axes being supplied by the transmission channels of the same integrated circuit, the other three sub-networks being supplied by the channels of the other integrated circuit.
    Said radar operates for example in millimeter waves, it is for example able to equip a motor vehicle.
    Other characteristics and advantages of the invention will become apparent with the aid of the description which follows, made with reference to the appended drawings which represent:
    - Figure 1, an illustration of the principle of realization of a network antenna used in a radar according to the invention;
    - Figure 2, an exemplary embodiment of the antenna where the number of lines is odd and where the number of columns is even;
    - Figure 3, an embodiment of the antenna bounded by a rectangular perimeter;
    - Figure 4, another embodiment of the antenna with three lines and five columns;
    - Figure 5, an example of transmission and reception system adapted to be coupled to said network antenna;
    - Figure 6, an example of antenna configuration suitable for a type of transmission and reception system composed of a given number of integrated circuits;
    - Figure 7, an example of antenna configuration suitable for a transmission system composed of two integrated circuits each comprising three transmission channels, four reception channels and a waveform generator;
    - Figure 8, an example of a first coding of the transmitted waves;
    - Figure 9, the array antenna of Figure 7 separated into two halves, left and right, symmetrical with respect to the intersection of the vertical and horizontal axes of the antenna;
    - Figure 10, examples of coding by phase modulation from ramp to ramp, according to rows or columns of the network antenna;
    - Figure 11, an example of a transmission system capable of being produced on a single integrated circuit;
    - Figure 12, the spectral bands respectively of the received signals demodulated by a first waveform for the signals from remission of said first waveform and a second waveform;
    - Figure 13, the filter banks arranged inside the two previous subbands;
    - Figure 14, the shape of the sums of a first formation of monopulse beam two planes and of a second formation of monopulse beam two planes;
    - Figure 15, the paces of the two ways differences of a first formation of monopulse beam two planes and of a second formation of monopulse beam two planes;
    - Figure 16, straight-line deviations obtained by using the aforementioned sums and differences channels.
    FIG. 1 illustrates the principle of production of a network antenna used in a radar according to the invention.
    In a radar according to the invention, there are available on the entire surface of the available antenna 1, adjacent antenna sub-networks operating exclusively in transmission 11 or reception 11, so as to form a two-plane monopulse beam formation in transmission and reception by dividing the antenna into four quadrants 1, 2, 3, 4 on transmission and reception, the monopulse beam formation in transmission being obtained by double coding, respectively in frequency 101, 102 and in phase 103, 104, according to the four quadrants of the antenna.
    The invention advantageously solves the problem posed by improving the resolution and the accuracy of angular localization of the antenna for a given antenna surface thanks to the multiplication of sum and difference diagrams produced both in transmission and in reception. The secondary and ambiguous lobes are limited due to the adjacency of the sub-networks and their uniform distribution.
    The range budget is optimized by the radiating surface of the antenna which is maximum, and by the fact that the emission and the reception are separated, which reduces the coupling, therefore the noise in reception
    The beam formation carried out in emission is limited to four beams obtained by summation and by differentiation which does not require significant computing resources
    The angular estimates obtained in azimuth and in elevation are obtained independently and these estimates are uncorrelated with each other.
    Composite antenna patterns are symmetrical in azimuth and elevation, ensuring consistent quality of detection and localization in the viewing area.
    It is possible to form wide field or narrow field diagrams simultaneously, for example for short-range and long-range detection.
    It is possible to adjust the level of the secondary lobes by adjusting the amplitude of the signals on transmission or reception on the various sub-networks.
    There is no switching device in the antenna, which is favorable for the range budget. The treatment is simple and easy to implement.
    FIG. 1 illustrates the principle of arrangement of the sub-networks 11, 12 on the surface of the antenna 10. Subsequently, the transmission sub-networks will be noted TX and RX the receiving sub-networks. Each sub-network is made up of a given number of radiant elements. An antenna of a radar according to the invention comprises P transmitting TX antenna sub-networks and Q receiving antenna RX sub-networks, the TX and RX sub-networks being adjacent to each other. They have for example an identical opening and adjusted to the desired coverage area. Preferably, all of the TX transmission subnetworks are identical and all of the RX reception subnetworks are identical. They are arranged to meet the following conditions:
    - Distribute on the antenna surface all of the TX, RX sub-arrays along several horizontal lines and several vertical columns to obtain a symmetrical distribution of these TX, RX sub-arrays along the vertical axis 5 and the horizontal axis 6 , these two axes passing through the geometric center of the antenna. In other words, each transmitting subnetwork 11 has a symmetrical transmitting subnetwork 11 ’with respect to the vertical axis and a transmitting subnetwork 11 ″ symmetrical with respect to the horizontal axis. Likewise, each reception sub-network 12 has a symmetrical reception sub-network 12 ’with respect to the vertical axis and a reception sub-network 12” symmetrical with respect to the horizontal axis;
    - Distribute the TX transmission sub-networks on the different lines and on the different columns so that at least two symmetrical sub-networks 11, 11 ′ with respect to the vertical axis are separated by the greatest possible horizontal distance and that at least two other symmetrical sub-networks 11, 11 ”with respect to the horizontal axis are distant from the greatest vertical distance possible between sub-networks, taking into account the limits imposed by the available surface of the antenna;
    - Distribute the RX receiving sub-networks on the different lines and on the different columns so that at least two symmetrical sub-networks 14, 14 'with respect to the vertical axis are distant from the greatest horizontal distance possible and that 'at least two other symmetrical sub-networks 12, 12 ”with respect to the horizontal axis are distant from the greatest possible vertical distance between sub-networks, taking into account the limits imposed by the available surface of the antenna;
    FIG. 1 shows an exemplary embodiment in the case of a rectangular surface, meeting the above conditions, where the number of rows is par, equal to 4, and the number of columns is even, equal to 6, and where P = Q = 12.
    Figure 2 shows an example of an embodiment where the number of rows is odd, equal to 3, and where the number of columns is even, equal to 4, and where P = Q = 6. In this case, the axis of symmetry horizontal 6 passes through TX, RX subnets which are themselves symmetrical with respect to this axis.
    Other configurations are of course possible, for example with a number P of transmitting antenna sub-networks different from the number Q of receiving antenna sub-networks, in particular according to the technological constraints imposed by the internal architectures of the integrated components used in transmission and reception. The outline 9 of the antenna may also not be rectangular as will be shown below in other example embodiments.
    In cooperation with a type of antenna as illustrated in FIGS. 1 and 2, a radar according to the invention:
    - Performs from a common oscillator reference a double coding of the radar transmission, by lines and by columns of transmission subnets according to two different types of coding, respectively by frequency modulation (according to columns 101, 102 of quadrants for example) and by phase modulation (according to lines 103, 104 of quadrants for example), so as to carry out orthogonal emissions both between the different lines or grouping of lines of subnetworks and between the different columns or grouping of subnetwork columns;
    - Forms a sum channel and a difference channel on transmission according to the azimuth angle by carrying out on transmission a grouping of part of the sub-networks of the left half 7 of the antenna and symmetrically a grouping identical with sub-networks of the right-hand part 8 of the antenna;
    - Forms a sum channel and a difference channel on transmission according to the elevation angle by carrying out on transmission a grouping of part of the sub-networks of the upper half 17 of the antenna and symmetrically a grouping identical to part of the sub-networks of the lower half 18 of the antenna
    - Forms a sum channel and a difference channel in reception according to the azimuth angle using the signals received from a first grouping of receiving sub-networks on the left half 7 and from a second identical and symmetrical grouping on the half right 8 of the antenna;
    - Forms a sum channel and a difference channel in reception along the elevation axis using the signals received from a first grouping of receiving subnets on the upper half 17 and from a second identical and symmetrical grouping on the half lower 18 of the antenna;
    - Forms the beams of composite radiation emission reception corresponding to the sum and difference channels, separately according to the azimuth and elevation angles;
    - Detects and estimates by target measurement the position of the targets from the sum and difference channels thus formed.
    These processing phases are described in more detail below using examples of antenna construction of a radar according to the invention.
    Before presenting these exemplary embodiments, we return to the principle of construction of such an antenna.
    Figures 3 and 4 illustrate the principle of embodiment of the antenna, the latter comprising three rows and five columns of antenna sub-arrays in the example presented.
    A first stage of construction consists in optimizing the number and the distribution of the antenna sub-networks of emission and reception on the surface of the antenna, surface which is limited.
    For this, we consider a planar array antenna whose outline is inscribed in a perimeter of determined dimensions. In the example of FIG. 3, the perimeter is a rectangle of horizontal length L and vertical length H, and whose angular coverage domain is imposed and corresponds to an opening cone at 3dB Δθ β ι radians in elevation and ΔΘ 3Ζ radians in azimuth. This antenna is made up of a certain number of sub-networks assigned to the transmission, and a certain number of sub-networks assigned to the reception.
    Conventionally, the maximum dimensions of the antenna transmission and reception sub-networks constituting this antenna are determined, ie a maximum vertical height h of the order of λ / Δθ β ι and a maximum horizontal width l = λ / Δθ 3Ζ .
    We choose to size the antenna sub-networks according to their maximum dimension to exactly cover the desired angular range, in order to restrict the number of radar transmission and reception channels. This leads to a horizontal dimension l = λ / Δθ β ι and to a vertical dimension h = λ / Δθ θ ι. We also choose to use the maximum radiation area in the dimensions allowed for the antenna, to optimize both the angular resolution and the range budget of the radar.
    In this case, an advantageous solution is to superimpose along the vertical axis P sub-network lines and align according to the horizontal plane Q sub-network columns, where P is the integer value of (H / h) and Q is the integer value of (L / Z).
    Typically, for an automotive radar operating at a frequency of 76 GHz, the wavelength is 3.9 mm the desired angular opening is for example of the order of 0.15 rd in elevation and 0.25 rd in Azimuth. The sub-networks are produced in “patch” type printed circuit technology, and the height of a sub-network is for example in this case of the order of 2.5 cm and its width of the order of 1.5 cm.
    Furthermore, the maximum dimensions of the antenna are even imposed, typically less than 8 cm in height and in width.
    Thus, according to these values, it is theoretically possible to set up 15 sub-networks 2.5 cm high and 1.5 cm wide along three lines and five columns in accordance with Figure 3.
    Then choose the number of TX subnets assigned to the transmission and the number of RX subnets assigned to the reception. The maximum number of antenna sub-networks having been determined, it is first of all necessary that the distribution of the transmission sub-networks and of the reception sub-networks is symmetrical both with respect to a horizontal axis 6 located at mid-height of the 'antenna, and relative to a vertical axis 5 located mid-width of the antenna.
    Secondly, at least two transmission sub-networks 41, 41 'and two reception sub-networks 43, 43' are symmetrical with respect to the vertical axis of symmetry so that the phase centers of each of these two sub-networks are distant by the greatest horizontal distance possible, taking into account the available installation width.
    Thirdly, at least two transmission subnets 42, 42 'and two reception subnets 43, 43' are symmetrical with respect to the horizontal axis of symmetry in such a way that the phase centers of each of these two subnetworks is distant by the greatest vertical distance possible, taking into account the available layout width.
    Fourth, the most uniform possible distribution of the transmission sub-networks and of the reception sub-networks is imposed on the antenna, the TX transmission and RX reception sub-networks being adjacent, and finally alternating.
    It is also necessary to minimize the number of transmission channels to simplify the physical architecture, reduce consumption and the risks of transmission-reception coupling.
    In this case, according to the previous example of an antenna comprising three rows and five columns of subnets, one arrives for example at the configuration example of FIG. 4 below, in which the TX and RX subnets are alternated along the vertical axis and along the horizontal axis. In the case of an antenna comprising an odd number of lines and / or columns, the construction symmetry requires that the axes of symmetry 5, 6 pass through the sub-arrays of the line and / or of the middle column, these being themselves symmetrical with respect to axis 5, 6.
    In the example of FIG. 4, the antenna comprises seven TX sub-networks and eight RX sub-networks and corresponds to an optimum case from the point of view of the radiation of the antenna, taking into account the available surface.
    However, with regard to automotive radars, the choice of the number of TX transmission subnetworks and the number of RX reception subnetworks may be constrained by the architecture of the associated transmission and reception microwave integrated circuits. to these subnets. The embodiments presented below will be adapted to architectures of imposed microwave integrated circuits.
    FIG. 5 shows an example of a transmission and reception circuit capable of being coupled to the sub-networks of an antenna 1 of a radar according to the invention. The circuit functions can be integrated on one or more semiconductor components.
    For the program, this circuit includes:
    - A voltage controlled oscillator 50, also called VCO (Voltage Controlled Oscillator) acting as a FMCW waveform generator, capable of supplying several emission channels;
    - A transmitter 51, composed of several transmission channels 511, 512, 51 i each comprising at least one power amplifier and a phase modulator with two states (0, π), each of these channels supplying an antenna sub-network TX-ι, TX 2 , TX ,.
    For reception, it includes:
    - The same VCO 50 to demodulate the signals received on the reception channels;
    - A receiver 52, composed of several reception channels 521, 522, 52i comprising at least one low noise amplifier, a synchronous demodulation function and a filter, each of these channels receiving a signal from an antenna reception subnet RXi, RX 2 , RXj.
    The reception channels carry out for example the direct demodulation of the different signals received by the signal from the VCO. The digital conversion of the received signals can also be integrated into the same component as the reception function.
    The VCO, TXi transmission and RXj reception functions can be integrated on different chips or on the same chip.
    Several levels of integration of transmission and reception circuits are possible. The antenna architecture can then advantageously be adapted to one or other of these integration levels.
    FIG. 6 presents an antenna configuration corresponding to a level of integration where two transmission channels are integrated on the same chip, three reception channels are integrated on the same chip with a separate VCO. Referring to FIG. 5, this amounts to having ΤΧί and TX 2 on the same chip, and RXi, RX 2 and RX3 on another chip. By using these integrated components, available in particular on the market, the antenna configuration comprises a multiple of 2 TX (2 transmission sub-networks) and 3 RX (3 reception sub-networks). In the case of FIG. 6, the multiple is 3, the antenna comprising 15 antenna sub-networks, including 6 TX transmission sub-networks and 9 RX reception sub-networks, these being arranged according to the rules defined above. Three transmission chips TX1, TX 2 and three reception chips RX1, RX 2 , RX3 are therefore used.
    FIG. 7 presents an antenna configuration corresponding to an integration level where three transmission channels, four reception channels and the VCO are on the same chip. With reference to FIG. 5, an integrated circuit therefore comprises 3 TX, 4 RX and a VCO, that is TX1, TX 2 and TX3 and RX-ι, RX 2 , RX3 and RX4 and the VCO on the same chip. Using this integrated circuit, the antenna configuration comprises a multiple of 3 TX and 4 RX. In the case of FIG. 7, two integrated circuits are used, the multiple being equal to 2. The antenna therefore comprises 14 antenna sub-networks, divided into 6 TX sub-networks and 8 RX sub-networks.
    To establish the configuration, an RX reception subnet is deleted with respect to that of FIG. 6. This is deleted on the middle column, the two remaining subnets being offset so as to be symmetrical with respect to the horizontal axis of symmetry 6.
    The configuration of FIG. 7 is an advantageous solution adapted to the use of 3 TX, 4 RX integrated circuits.
    Subsequently we will consider, by way of example, this solution as a reference solution.
    FIG. 8 shows a first type of coding used by the invention, more precisely the frequency coding applied according to one dimension of the antenna, the coding applied according to the other dimension being the phase coding which will be presented below.
    According to the invention, this first coding of the transmitted wave is carried out by generating sequences of frequency ramps 81, 82 which are identical but offset in frequency. Each of these ramps is generated by a VCO.
    Coding can be carried out preferentially by generating a first emission waveform 81 periodically producing a first frequency ramp using a first VCO, denoted VCOa, and a second waveform 82 identical and synchronous with the first but offset in frequency by a difference ôf, using a second VCO, denoted VCOb, the two VCOs being controlled by the same reference clock. These frequency ramps are coherent and orthogonal to each other.
    Such coding is known. It is notably described for example in the article by Matthias Steinhauer "Millimeter-Wave-Radar Sensor Based on a Tranceiver Array for Automotive Applications", IEEE Transactions on Microwave Theory and Techniques (Volume 56, pages 262-269, February 2008).
    We refer to Figure 9 which shows the antenna shown in Figure 7. This is separated into two parts, two halves. A first half 91 comprises the antenna sub-networks located on the left side of the antenna as well as the upper RX7 reception sub-network of the middle column. The second half 92 comprises the antenna sub-networks located on the right side of the antenna as well as the lower RX8 reception sub-network of the middle column. The coding illustrated in FIG. 8 makes it possible to distinguish these two halves.
    Thus, according to the invention, the waveform originating from the VCOa feeds the TX transmission channels of a first half 91 and is also used as a demodulation reference for the RX reception channels of this first half 91.
    Likewise, the waveform coming from the VCOb feeds the TX transmission channels of the other half 92 and is also used as demodulation reference for the RX reception channels of this other half 92.
    For example, the VCOa feeds the left side of the antenna and the VCOb feeds the right side symmetrically.
    By numbering the different antenna networks TX and RX according to FIG. 9, the VCOa is for example associated with the subnets TX + TX2, TX 3 for the transmission and with the subnetworks RX1, RX2, RX3, RX7 for the demodulation on reception . Symmetrically, the VCOb is associated with TX4, TX 5 , ΤΧθ for transmission and RX 4 , RX5, RX6, RXs for demodulation on reception.
    FIG. 10 illustrates the second coding, carried out according to the other dimension, that is to say along the lines. This second coding performed this time in phase, is carried out at the level of the various transmission channels 511, 512, 51 i of the transmission and reception circuit.
    This coding is carried out from ramp to ramp, using a two-phase modulation, comprising two possible phase states, 0 and π. The different codes are orthogonal to each other. Preferably, the different transmission channels of the same line are modulated by the same code.
    Preferably, the codes chosen are Hadamard codes. These codes include M = 2 P moments and are perfectly orthogonal to each other on a burst of M successive ramps. Other orthogonal codes are of course possible.
    FIG. 10 presents three phase codes 1001, 1002, 1003 carried out from ramp to ramp, that is to say that the phase code is liable to vary from one ramp to the next. The first code 1001 assigns the phase value 0 on all the ramps. The second code 1002 assigns the value 0 every other ramp alternately with the value π. The third code assigns code 0 on two successive ramps then the π code on the following two ramps and so on.
    Referring to FIG. 9, the first phase code 1001 is assigned to the first line, ie to the transmission channels supplying the TX1 and TX4 subnets. The second phase code 1002 is assigned to the first line, ie to the transmission channels supplying the TX2 and TX5 subnets. The third phase code 1003 is assigned to the first line, namely to the transmission channels supplying the TX3 and TX6 subnets.
    FIG. 11 shows the diagram of the transmission and reception circuit assigned to a half-antenna, more particularly to the left part 91, all the elements of the circuit being able advantageously to be integrated in the same component. Compared to the diagram in FIG. 5, this presents the three transmission channels 511, 512, 513 and the four reception channels 521, 522, 523, 524. The VCO delivers the low-level transmission signal according to the first frequency ramp 81 thus carrying out the first code, in frequency. The signal is amplified by a power amplifier 111 in each transmission channel, the latter is followed by a phase shifter 112. The latter applies the code associated with the transmission channel as described above.
    On reception, each channel has at its input a mixer 114 receiving on a first input the signal received from the associated subnetwork, possibly amplified by a first amplifier 113. The second input of the mixer receives the ramp signal 81 supplied by the VCO. The received signal is thus demodulated and amplified at the output of the mixer by a low noise amplifier 115. Filtering 116 and analog-to-digital conversion follow this amplifier, the reception channel delivering as output a digitized signal capable of being processed by the processing means of the radar. Symmetrically, the same circuit is coupled to the right part 92 of the antenna, with a VCO delivering the offset frequency ramp 82.
    In the processing at the output of the reception chain, from these received and digitized signals, frequency separation and distance compression of the signals originating from the emission produced by the two VCOs are carried out.
    In an FMCW radar, it is known to a person skilled in the art to limit the output bandwidth of the receiver using a low pass filter whose cutoff frequency is adjusted to the maximum instrumented range, so to process only useful signals.
    Thus for an AF modulation band and a ramp duration T, an echo corresponding to a target located at the maximum instrumented distance D max from the radar, the maximum beat frequency fbmax is (excluding doppler which introduces a negligible difference):
    r _ Max 2D AF, ..
    Jbmax ~ T c 1 / where C is the speed of light.
    In the usual case where the emission signal is used to demodulate the reception signal, the frequency spectrum resulting from the synchronous demodulation thus extends from 0 to + fbmax
    The cutoff frequency of the low pass filter is thus chosen to be equal to f bmax .
    According to the particular architecture illustrated in FIG. 9, the different receivers receive from the same target echoes originating from the emission originating from the VCOa in a band comprised between F and F + AF and other echoes originating from the emission from VCOb in an F-ôf and F-5f + AF band.
    For receivers whose demodulation signal is generated by the VCOa, the frequency band in reception after demodulation thus extends from 0 to f bmax for the signals transmitted from the VCOa and extends from -ôf to -ài + f bmax for signals transmitted from the VCOb.
    Thus, by choosing a frequency difference ôf between the two VCOs greater than the beat frequency f bmax , the frequency spectra of the reception signals demodulated by one of the VCOs and originating from the emissions from the two VCOs occupy separate bands. and can be separated by filtering, this is illustrated in Figure 12.
    FIG. 12 illustrates the spectral bands 121, 122 corresponding to the signals received demodulated by the VCOa, VCO of the first transmission and reception circuit, in a system of axes where the abscissas represent the frequencies f and the ordinates the amplitude of the spectrum. The first domain 121 comprised between 0 and f bmax is the spectral domain of the echoes resulting from the emission resulting from VCOa after demodulation by VCOa- The second domain 122 comprised between 5f and tiï + f bmax is the spectral domain of the echoes resulting from l 'emission from VCO B after demodulation by VCOa Symmetrically with respect to the axis f = 0, the domains 121' and 122 'respectively represent the spectral domain of the echoes resulting from the emission from VCOa after demodulation by VCOb and the spectral domain echoes resulting from the emission from VCOb after demodulation by VCOb.
    Thus by performing bandpass filtering on reception comprising two distinct sub-bands as illustrated in FIG. 12, it is possible to separate on each reception channel the signals resulting from the transmissions of the two VCOs. In other words, frequency coding makes it possible to separate the signals according to whether they are transmitted by the left part 91 or the right part 92 of the antenna.
    Assuming the phase at the origin of the VCOa frequency ramps equal to zero, the signal received at time t on the receiver of index j associated with the sub-network RXj, after demodulation by the VCOa and relative to a transmission performed from the VCOa supplying the transmitter of index i associated with TXj, is written:
    (D iW + DjÇfl
    Ae 2} c F / (D ^ + DjiJpAF e ~ 2 ”(-ci-2v r t
    A) τ e-2jn φΟ ^ τηΤΓ) (2) where the exponent of the first term of the product is the phase term which is a function of the distance and the angle of the target with respect to TXi and RXj, and the exponent of the second term is the frequency f b , frequency ambiguous distance distance / Doppler.
    with
    A: amplitude of the received signal
    AF: modulation band of the FMCW ramp
    F: VCOa ramp start frequency
    Dj (t): distance between the phase center of the TX transmission subnetwork, and the target at time t
    Dj (t): distance between the phase center of the RX receiving subnetwork, and the target at time t vr: speed of movement of the target t: time, with t = mTr + ^ Pt where t = mTr + 2C: speed λ: wavelength of the emission signal considered as constant relative to the doppler effect (pQiÇmTr): phase at the origin of the frequency ramp emitted by the transmitter TXj at the recurrence Tr of rank m , depending on the phase code applied to the TX,
    VS
    The second term of equation (2) can be simplified by taking into account the fact that D, (t) is substantially equal Dj (t) and by posing:
    D (t) =
    P - 2 ^ (3) (4)
    CT
    On the scale of a frequency ramp 81, 82, the distance from the target can be considered as constant, and D (t) = D. In that case :
    and f b =
    2DÙF
    CT + F d (5) (Dgn + Djff)
    Si'jit) = Ae _2; 7r · -c- F. e-TUfb t e -2jncpOAmTr) (β)
    Similarly, assuming the phase at the origin of the VCOb frequency ramps equal to zero, the signal s kJ (t) received on the receiver of index j associated with RXj is written, after demodulation by the VCOa relative to a transmission carried out from the VCOb at the frequency F + ôf supplying the transmitter of index k associated with TX k , is written:
    (D fc (t) + D, - (t))
    S k j (t) = Ae ~ 2in '-C- ( p + <V) e -2yrc (y 6 + <S /) t e -2yrcpO fc (nirr) (jj where:
    - D k is the distance between the phase center of the TX sub-network TX k and the target;
    - <pO fc (mTr) is the phase at the origin of the frequency ramp emitted by the transmitter TX k at the recurrence Tr of rank m, according to the phase code applied to TX k .
    For receivers which the demodulation signal is generated by the VCoA, receiving frequency band after demodulation thus extends from 0 to P FBMAX ° ur the signals emitted from the VCoA and extends to -of -6i + f bmax for signals transmitted from the VCOb. (neglecting the Doppler frequency which is very small compared to the deviation of the transmission frequency f).
    It is then possible to separate on reception the signals coming from the two VCOs, by bandpass filtering, preferably carried out in digital, typically by a Fourrier transform (FFT or DFT)
    The demodulation followed by the Fourrier transform conventionally corresponds to the compression of the signal in distance, according to a resolution AD =
    2 F
    At the output of the filtering, the signal is broken down into N distance filters (or distance boxes) according to FIG. 13.
    This breakdown is carried out identically for the signals emitted by the VCOa and demodulated by this same VCOa and for the signals emitted by the VCOb and demodulated by the VCOa. Figure 13 therefore illustrates the filter banks thus created arranged inside the two sub-bands 121, 122. A first bank 131 of N distance filters corresponds to the emission from VCOa and to the demodulation by VCOa. A second bank 132 of N distance filters corresponds to the emission from VCO B and to the demodulation by VCO A. It is thus possible to decompose the two sub-bands 121, 132 into distance boxes.
    Thus, in the first bank 131 of N filters corresponding to the transmission TX, coming from VCO A , for the receiver RXj, the phase of the signal at the output of a distance filter 31 of rank n at the recurrence m is equal, d 'after relation (6), to:
    <Pi 7 (m R) = —2jn. ~ ---- F - 2jn (pOfmTr) (8)
    In the second bank 132 of N filters corresponding to the transmission TX k from the VCO A , for the receiver RXj, the phase of the signal at the output of a distance filter 32 of rank n at the recurrence m is equal, according to relation (7), to:
    (mTr) = —2dn. ( Pfe (t) + p J (t) ) (p + δ β _ 2 jn (pQ k (mTr) (9)
    The signals output from these filters of rank n with the recurrence of order m can be written in simplified form respectively:
    (DiOT + D / t))
    UijÇn, m) = Ae ~ 2} n '-c- F e ~ 2 ^ <M mTr î (10) and
    U k> j (n, m) = Ae ~ 2jn --C- ( F + ^ n e -2jn <pO k (mTr) (11 )
    After separation in distance by the previous processing, corresponding to the short time, carried out on each frequency ramp, the radar processing means carry out for example in a conventional manner a coherent integration processing on the doppler axis, aiming to optimize the ratio signal to noise and to separate the targets according to their speed, by digital Fourier transform (FFT or DFT).
    This processing is preceded by a phase correlation aimed at separating the signals received corresponding to the different TX transmission lines from the antenna.
    This processing is carried out on a set of M successive frequency ramps, for each distance filter or possibly on a limited number of distance filters corresponding to the desired detection domain. In expressions (10) and (11) above, the distance terms DftfDjÇt ^ and D k (t) are a function of both the initial distance, the radial speed and the angular location of the targets at during the Doppler burst.
    Thus, for a given target, taking as origin O the physical center of the network antenna and noting:
    . cbcg height of the phase center of the antenna sub-network RX, relative to O. dyy. horizontal distance from the phase center of the RX antenna subarray, with respect to O. dx k height of the phase center of the antenna sub-network T k with respect to O dy k horizontal distance from the phase center of the antenna sub-network TX k with respect to O. dxk '. height of the phase center of the antenna sub-network T k with respect to O. dyc. horizontal distance from the phase center of the antenna sub-network TX k with respect to O
    - θ αζ : azimuth angle of the target considered
    - O e i: elevation angle of the target considered
    - D o : Distance between the antenna and the target at the time origin of a Doppler burst made up of M successive ramps
    We can write :
    D k (t) = D o + V r t + (dx fc sin (0 eZ ) + dy k sin (6 az )) Dy (t) = D o + V r t + (dxj sin (0 eÎ ) + dyj sin (θ αζ )) Dflt) = D 0 + V r t + (dxt sin (0 and ) + dy ( sin (0 az ))
    The signals U it j (n, m) at the output of the distance filtering can thus be written according to the following relation (12):
    -iinDnF. ( dx i + dX /) sin (e ei ) + (dy t + dy,) sin (6 az )
    Uifn.m) = Ae-pr-eV ^ e- 21 ”· -cp e -2jn where the exhibitors successively represent:
    - A phase term depending on the distance;
    - A phase term depending on the speed of the target;
    - A phase term depending on the angular location;
    - A phase term depending on the phase at the origin of the frequency ramp m for the emission channel associated with the TXi subnetwork, according to the phase code applied.
    and:
    U kJ (n, m) =
    (dx fc + rf% y) sin (e eÎ ) + (dy fe + dyy) sin (e az ) ff + <V) e -2y'7r <pO fe (7n7y) (.jg) with
    2v r F „2v r (F + 5 /) —— # (14)
    The phase correlation operations and the Doppler compression are carried out in a single operation by performing a Fourier transform on the signal at the output of the distance filter modulated by the conjugate of the phase code applied to the TXi considered:
    <M-1 m = 0
    Uij (n, m) β +2; π e + 2pr lm
    M (15)
    For a doppler frequency target F d = - corresponding to the center of a doppler filter of rank /, the output of the doppler filter corresponding to the receiver RXj for the signal emitted by the transmitter TX, can be written in simplified form:
    M4, y (n, /) = Ae c e
    c (16) and similarly, the output of the doppler filter corresponding to the receiver RXj for the signal transmitted by the transmitter TXk can be written in simplified form;
    -4jnD 0 (F + SP> ( dx k + d% ; ) sin (e eÎ ) + (dy fc + dyj) sin (. <3 az ·)
    W kJ (n, l) = Ae c e ~ 2} n 'c • (.F + sf) (i )
    Thus, by repeating the configuration of FIG. 9, it is possible to calculate for each receiver the response corresponding to each transmitter in a separate manner.
    For example, for the reception channel associated with the RXi subnet, the responses are as follows for the different transmissions:
    -4jnD 0 F „, _ (dx 1 + dx 1 ) sin (e e i) + (dy 1 + dy 1 ) sin (e az ) r ,
    TXi: Γ) = Ae ~ ^ e ~ 2jn - c F
    -4] nD 0 (F4Sf) „(dx 4 + dxi) sin (e eZ ) + (dy 4 + dyi) sin (0, az ) tr ,,
    TX 4 : W 4il (n, Z) = Ae-c- e ~ 2] π '-c- ( F + W
    -4jnD 0 F „(dx 2 + dxi) sin (e eZ ) + (dy 2 + dyi) sin (0 az )
    TX 2 : W 2/1 (n, Z) = Ae ~ - e ~ 2j7t · -c F —4jnDo (F + Sf) (dx 5 + dxi) sin (e e j) + (dy 5 + dyi) sin (0 az )
    TX 5 : W 5Λ ( n , Z) = Ae-c- e ~ 2jn · -c- (F + 5 /)
    -4jnD 0 F „(dx 3 + dxi) sin (e e ;) + (dy 3 + dyi) sin (0 az ) r ,
    TX 3 : Ws ,! (N, Z) = Ae ~ ^ e ~ 2jn · -c F
    -4jnD 0 (F + Sn „._ (dX6 + dxi) sin (e e ;) + (dy 6 + dyi) sin (e az ) / r ,,
    TX 6 : W 6> 1 (n, Z) = Ae-c- e ~ 2} π · -cil the same goes for the signals received on the other reception sub-networks, which makes it possible to calculate from a generally for each range of rank n and for each doppler filter of rank Z, the responses W pq (Z, n) for p = 1 to 6 (index of the transmission sub-network) and for q = 1 to 8 (index of the receiving sub-network).
    After having separated the signals received thanks to the two types of codes used, it is by summing and differentiating these responses W p> q (l, ri) that the sum and difference channels are subsequently formed.
    The following describes the formation of the sum and difference channels on transmission, then on reception by the radar.
    From the previous processing, the signals received W pq (l, n) on each receiver of index q are separated according to their original transmitter of index p. As previously, the reception channel associated with the reception sub-network RX is called the RX receiver. Likewise, the reception channel associated with the TX transmission subnetwork is called TX transmitter.
    These signals being thus separated, the sum Σ and difference Δ channels for transmission are then formed separately and for each receiver of index p:
    in azimuth:
    Xeaz q (n, Z) = Σρ = ^ p, q (n, l) Aeaz q (n, l) = Σρ = ι Wp.qCn, l) - Σ (no (18) (19) and in elevation:
    Seel q (n, l) = Zeaz q (n, l) = Σρ = %, < 0τ0 (20)
    It is noted in this example, that for the elevation path, only four transmitters are used in the vertical plane, the transmitters of the central line being unable to contribute to the realization of the difference in elevation path. In particular, relation (19) translates the fact that the two transmission subnets TX2 and TX5 of the middle line are not used because they cancel each other out and therefore do not provide information.
    The processing focuses the emission signal on three beams. For a given axis, azimuth or elevation, the sum channel and difference channel antenna diagrams thus formed are identical in amplitude for all the receivers and their phase differs according to the position of the different receivers in the antenna network.
    The beam forming processing on reception is carried out by associating the signals received from each receiver after the beam forming processing on transmission as described above.
    This treatment consists in independently carrying out the sum and difference channels along the two axes, azimuth and elevation.
    In azimuth:
    Zeraz (n, Γ) = Zeaz q (η, Z) (22)
    Aeraz (n, i) = Σ ^ ΖΪΔβαζ ^ η, Ι) -Y ^ Aeaz q (n, l) (23)
    Note in this example, that only six receivers are used in the horizontal plane to carry out the delta path in azimuth, the receivers of the central column being unable to contribute to this realization.
    In elevation:
    ZerelÇn, Γ) = Zeraz (n, Z) = Σ ^ = θ Zeaz q (η, Z) (24)
    Aerel (n, Z) = (Aeebfln, Z) + Aeel 4 (n, Z) + Aeel 7 (n, Z)) - (Aeel 3 (n, Z) +
    Aeel e (n, Z) + Aeel 6 (n, Z)) (25)
    We note in this example, that only six receivers are used in the vertical plane to carry out the delta path in elevation, the receivers of the central line being unable to contribute to this realization.
    This processing focuses on reception. The final result corresponds to the multiplication of transmission / reception diagrams.
    FIGS. 14 and 15 represent the appearance of the antenna diagrams obtained after channel formation in transmission and reception, corresponding to the example of antenna considered in FIG. 9, for the sum and difference channels, FIG. 14 illustrating the sum channel and FIG. 15 illustrating the two difference channels.
    These diagrams are obtained without amplitude weighting of the sub-networks. If necessary, the level of the secondary lobes can be further reduced by applying such weighting.
    Conventionally, the signals calculated at the output of the sum channel are used for the detection of targets. The localization of the targets is obtained by monopulse deviation, from the sum and difference channels, for example by forming to within a scale factor:
    ecartoaz (n, l) = sign (arg (rieraz (n, Z)) * arctg (and ecartoel (n, V) = stgne (arg (ziereZ (n, Z)) * arctg (i (26)
    FIG. 16 presents the straight-line deviations obtained by these treatments, still for the antenna configuration of FIG. 9. The top curve 161 represents the straight-line deviation in elevation. The bottom curve 162 represents the straight line in azimuth. The azimuth and elevation angles of the targets can thus be measured using these lines. The slopes of the latter are sufficiently steep to allow precise measurements. These lines also measure the elevation angle and the azimuth angle independently.
    The invention advantageously makes it possible to improve the resolution and the accuracy of angular localization of the antenna for a given antenna surface thanks to the multiplication of sum and difference diagrams produced both in transmission and in reception. The invention also has the advantages mentioned below.
    The secondary and ambiguous lobes are limited due to the adjacency of the sub-networks and their uniform distribution.
    The range budget is optimized by the radiating surface of the antenna which is maximum, and by the fact that the emission and the reception are separated, which reduces the coupling, therefore the noise in reception.
    The beam formation performed in transmission is limited to four beams obtained by summation and by differentiation which does not require significant computing resources.
    The angular estimates obtained in azimuth and in elevation are obtained independently and these estimates are uncorrelated with each other.
    Composite antenna patterns are symmetrical in azimuth and elevation, ensuring consistent quality of detection and localization in the viewing area.
    It is possible to form wide field or narrow field diagrams simultaneously, for example for short-range and long-range detection.
    It is possible to adjust the level of the secondary lobes by adjusting the amplitude of the signals on transmission or reception on the various sub-networks.
    There is no switching device in the antenna, which is favorable for the range budget;
    Finally, the treatment is simple and easy to implement.
    The invention has been presented for codings of the transmitted wave, in frequencies and in phases, carried out according to lines or columns, according to left and right parts, it is of course possible to carry out these codings according to other sub - sets of transmission and reception sub-arrays when these make it possible to discriminate parts of antennas.
    权利要求:
    Claims (18)
    [1" id="c-fr-0001]
    1. Radar with continuous wave and frequency modulation according to frequency ramps, called FMCW, comprising at least one network antenna (10) composed of transmission sub-networks and reception sub-networks, a transmission system and reception and processing means, characterized in that:
    - the distribution of the transmission sub-networks and the reception sub-networks is symmetrical both with respect to a vertical axis (5) and with respect to a horizontal axis (6);
    - at least two emission sub-networks (11, 1 T, 41.41 ’) symmetrical with respect to said vertical axis (5) are separated by the greatest possible distance;
    - at least two transmission sub-networks (11, 11 ”, 42, 42”) symmetrical with respect to said horizontal axis (6) are distant from the greatest possible distance;
    - at least two receiving sub-networks (12, 12 ’, 43, 43’) symmetrical with respect to said vertical axis (5) are distant from the greatest possible distance;
    - at least two receiving sub-networks (12, 12 ”, 43, 43”) symmetrical with respect to said horizontal axis (6) are distant from the greatest possible distance;
    - A first coding (101, 102) of the wave transmitted by said transmission subnets being carried out by frequency shift of said ramps (81, 82) between the different transmission subnetworks;
    - A second coding (103, 104) of the wave emitted by said transmission subnets being carried out by phase modulation from frequency ramp to frequency ramp between the different transmission subnetworks.
    [2" id="c-fr-0002]
    2. Radar according to claim 1, characterized in that in said first coding, a first half of the transmission sub-networks (7) is supplied by a first FMCW waveform and the second half (8) is supplied by the same waveform shifted in frequency, the two halves being symmetrical with respect to said vertical axis (5).
    [3" id="c-fr-0003]
    3. Radar according to claim 1, characterized in that in said first coding, a first left half of the transmission subnets (91) is fed by a first FMCW waveform and the second left half (92) is fed by the same frequency-shifted waveform, said two halves being symmetrical with respect to the intersection of said vertical axis (5) and said horizontal axis (6).
    [4" id="c-fr-0004]
    4. Radar according to any one of the preceding claims, characterized in that, in said second coding, the waves supplying the various transmission sub-networks belonging to the same line (1001, 1002, 1003), along the axis horizontal, are coded by the same phase code.
    [5" id="c-fr-0005]
    5. Radar according to claim 1, characterized in that, in said first coding, a first half of the transmission sub-networks is supplied by a first FMCW waveform and the second half is supplied by the same form of wave shifted in frequency, the two halves being symmetrical with respect to said horizontal axis (6).
    [6" id="c-fr-0006]
    6. Radar according to any one of the preceding claims, characterized in that, in said second coding, the waves feeding the different transmission sub-networks belonging to the same column, along the vertical axis, are coded by the same phase code.
    [7" id="c-fr-0007]
    7. Radar according to any one of the preceding claims, characterized in that said transmission and reception system comprises a first waveform generator (VCOa) generating a first FMCW waveform (81) and a second waveform generator (VCOb), synchronous and coherent with the first, generating the other waveform FMCW (82) shifted in frequency.
    [8" id="c-fr-0008]
    8. Radar according to claim 7, characterized in that each of said waveform generators (VCOa, VCOb) is used both for transmission (51) and for synchronous demodulation (114) of the reception signals ( 52).
    [9" id="c-fr-0009]
    9. Radar according to claim 8, characterized in that the frequency difference between said waveform generators (81, 82) is chosen so that the beat frequencies of the reception signals demodulated by a same form generator d 'wave (VCOa) occupy separate frequency bands (121, 122) depending on whether said received signals come from a transmission from the same generator (VCOa) or the other generator (VCOb).
    [10" id="c-fr-0010]
    10. Radar according to any one of the preceding claims, characterized in that the reception signals resulting from the various transmission sub-networks are separated by filtering (131, 132) and by reception correlation, respectively according to their frequency band (121 , 122) and according to their phase modulation code.
    [11" id="c-fr-0011]
    11. Radar according to any one of the preceding claims, characterized in that a first formation of monopulse beams two planes comprising a sum channel and two reception channels is carried out in transmission by said processing means, on each associated reception channel to a receiving subnetwork, using signals from all or part of the transmitting subnets.
    [12" id="c-fr-0012]
    12. Radar according to claim 11, characterized in that a second monopulse beam formation two planes comprising a sum channel and two difference channels is performed in reception by said processing means, by associating all or part of the signals received on all or part
    5 of the reception channels.
    [13" id="c-fr-0013]
    13. Radar according to claims 11 and 12, characterized in that the detection and the angular localization of targets are carried out on the basis of the signals resulting from the product of the monopulse beams of emission and the
    10 monopulse reception beams.
    [14" id="c-fr-0014]
    14. Radar according to any one of the preceding claims, characterized in that the phase modulation code applied is a two-phase code having the value 0 or π.
    [15" id="c-fr-0015]
    15. Radar according to any one of the preceding claims, characterized in that the phase modulation code applied is a Hadamard code.
    20
    [16" id="c-fr-0016]
    16. Radar according to any one of the preceding claims, characterized in that said antenna comprises six transmission sub-networks and eight reception sub-networks, said transmission and reception system comprising two integrated circuits each comprising a generator waveforms (50, VCOa, VCOb), three emission channels (511, 512, 513) and four
    25 reception channels (521, 522, 523, 524), the three transmission sub-networks arranged on one side of one of said axes (5, 6) being supplied by the transmission channels of the same circuit integrated, the other three subnetworks being supplied by the transmission channels of the other integrated circuit.
    [17" id="c-fr-0017]
    17. Radar according to any one of the preceding claims, characterized in that it operates in millimeter waves.
    [18" id="c-fr-0018]
    18. Radar according to any one of the preceding claims, 5 characterized in that it is able to equip a motor vehicle.
    1/11
    103
    104 =>
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    同族专利:
    公开号 | 公开日
    EP3315994A1|2018-05-02|
    US20180120427A1|2018-05-03|
    US10620305B2|2020-04-14|
    FR3058227B1|2018-11-02|
    EP3315994B1|2021-03-10|
    ES2870021T3|2021-10-26|
    引用文献:
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    EP1533866A1|2003-11-18|2005-05-25|Thales|Adaptive phased array antenna with digital beam forming|
    EP2312335A1|2009-10-12|2011-04-20|Thales|Radar with high angular precision, in particular for the function for detecting and avoiding obstacles|
    FR2987683A1|2012-03-02|2013-09-06|Thales Sa|RADAR WITH LOW PROBABILITY OF INTERCEPTION|
    JP3473563B2|2000-08-17|2003-12-08|日産自動車株式会社|Braking control device|
    TWI369621B|2008-10-03|2012-08-01|Ind Tech Res Inst|Yield evaluating apparatus and method thereof|
    CN102866387B|2012-10-16|2014-06-04|清华大学|Millimeter wave frequency modulated continuous wave two-unit phased array distance and velocity measurement monolithic radar transceiver|
    CN104871031B|2012-12-19|2017-05-24|索尼公司|A method for operating a handheld screening device and a handheld screening device|
    US9541639B2|2014-03-05|2017-01-10|Delphi Technologies, Inc.|MIMO antenna with elevation detection|
    JP6492377B2|2014-08-28|2019-04-03|日本無線株式会社|Orthogonal separation device and orthogonal separation method|DE102018117688A1|2017-08-18|2019-02-21|Infineon Technologies Ag|Radar frontend with RF oscillator monitoring|
    JP6844525B2|2017-12-26|2021-03-17|株式会社デンソー|Antenna device|
    DE102018116378A1|2018-07-06|2020-01-09|Valeo Schalter Und Sensoren Gmbh|Method for determining at least one object information of at least one target object that is detected with a radar system, in particular a vehicle, radar system and driver assistance system|
    JP2020051802A|2018-09-25|2020-04-02|パナソニックＩｐマネジメント株式会社|Radar device and target determination method|
    JP2020148754A|2019-03-07|2020-09-17|パナソニックＩｐマネジメント株式会社|Radar device|
    US11262434B2|2019-04-01|2022-03-01|GM Global Technology Operations LLC|Antenna array design and processing to eliminate false detections in a radar system|
    DE102019110525B4|2019-04-23|2021-07-29|Infineon Technologies Ag|CALIBRATING A RADAR SYSTEM|
    US11181614B2|2019-06-06|2021-11-23|GM Global Technology Operations LLC|Antenna array tilt and processing to eliminate false detections in a radar system|
    JP2021081282A|2019-11-18|2021-05-27|パナソニックＩｐマネジメント株式会社|Radar system|
    US20210173042A1|2019-12-09|2021-06-10|Nxp Usa, Inc.|Method and System for Frequency Offset Modulation Range Division MIMO Automotive Radar|
    法律状态:
    2017-09-29| PLFP| Fee payment|Year of fee payment: 2 |
    2018-05-04| PLSC| Publication of the preliminary search report|Effective date: 20180504 |
    2018-09-28| PLFP| Fee payment|Year of fee payment: 3 |
    2019-09-27| PLFP| Fee payment|Year of fee payment: 4 |
    2020-10-13| PLFP| Fee payment|Year of fee payment: 5 |
    2021-09-30| PLFP| Fee payment|Year of fee payment: 6 |
    优先权:
    申请号 | 申请日 | 专利标题
    FR1601552|2016-10-27|
    FR1601552A|FR3058227B1|2016-10-27|2016-10-27|FLEXW MULTIFUNCAL RADAR, IN PARTICULAR FOR AUTOMOBILE|FR1601552A| FR3058227B1|2016-10-27|2016-10-27|FLEXW MULTIFUNCAL RADAR, IN PARTICULAR FOR AUTOMOBILE|
    ES17197668T| ES2870021T3|2016-10-27|2017-10-23|FMCW multibeam radar, in particular for automobile|
    EP17197668.1A| EP3315994B1|2016-10-27|2017-10-23|Multibeam fmcw radar, in particular for a motor vehicle|
    US15/792,700| US10620305B2|2016-10-27|2017-10-24|Multibeam FMCW radar, in particular for automobile|
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