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    • 专利摘要:
    Controller and control method of a diode battery. Controller (200) and control method of a diode battery (100) capable of generating high power pulses and low pulse time thanks to a N-channel MOSFET transistor (400) connected to the diode stack (100) through of the drain (D) and at least one capacitor (C1) of high capacitance and low resistance equivalent series that is charged when the transistor (400) is open. In this way, the resonance frequency of the RLC circuit formed by the at least one capacitor (C1), the parasitic resistance (R) and the parasitic inductance (L) of the electric path is reduced, eliminating the pulse time limitations (tp) imposed by said RLC circuit. (Machine-translation by Google Translate, not legally binding)
    公开号:ES2710080A1
    申请号:ES201731228
    申请日:2017-10-18
    公开日:2019-04-22
    发明作者:Hervás Sergio Rodríguez；Rodas Miguel Sánchez；Rivera Horacio Lamela
    申请人:Universidad Carlos III de Madrid；
    IPC主号:
    专利说明:

    [0001]
    [0002] Controller and method of control of a diode battery
    [0003]
    [0004] Object of the invention
    [0005]
    [0006] The present invention relates to the field of electronics, and more specifically to the sector of the technique dedicated to laser source controllers.
    [0007]
    [0008] Background of the invention
    [0009]
    [0010] In recent years, biophotonics has concentrated a great scientific and business interest. Thanks to recent technological advances, more and more information can be extracted from the interaction between light and biological tissues, giving rise to more advanced and precise diagnostic techniques. In particular, the optoacoustic effect (also called photoacoustic) is a physical phenomenon based on the conversion of ultra-short pulses of light in pressure waves, propagated by the medium in the form of ultrasound. When a pulse of light is absorbed by a material, it produces a variation of the temperature that generates, in turn, a variation of pressure that propagates through the medium. This phenomenon can be used in biomedicine to perform Optoacoustic Tomography (OAT), benefiting from the main characteristics of optical imaging techniques (high contrast and the possibility of performing a spectroscopic analysis) and imaging techniques. by ultrasound (high resolution), because sound waves have less scattering in biological tissues than light. In addition, Optoacoustic Tomography uses a non-ionizing radiation, which greatly reduces the possibility of producing damage to these tissues. Therefore, it can be said that the OAT is a novel medicine technique for the diagnosis of cardiovascular diseases such as atherosclerosis or cancer.
    [0011]
    [0012] To be able to use a laser source in OAT applications, it must meet a series of requirements: pulses of light of high energy (several ^ J or even mJ), ultra short (tens or hundreds of ns) and repetition frequencies of the order of kHz to improve the resolution and the acquisition times of the measures. For this, traditionally, solid state lasers (Nd: YAG or Ti: Sapphire) or dye lasers have been used. However, in the last decades, high power laser diodes (HPLD) have been developed, which are cheaper, more compact and have a capacity of greater commutation, being able to increase the frequency of the system several orders of magnitude. However, the lack of power with respect to the other types of lasers makes the combination of multiple HPLDs necessary.
    [0013]
    [0014] In Figure 1 a possible strategy for the combination of multiple HPLDs through a diode stack (100, DLS, from the English 'diode laser stack') is presented schematically. Said diode stack (100) comprises a plurality of HPLD diodes (110) grouped into one-dimensional arrays called diode bars (120) (DLB). The batteries of diodes laser (or 'stacks of diodes') are used habitually in applications LIDAR (of the English 'Laser Imaging Detection and Ranging'), in which typically require frequencies of repetition of the order of 100 Hz and times of pulse relatively widths (of the order of hundreds of microseconds). However, the adaptation of this operating regime to the requirements of the OAT applications represents a significant technological challenge.
    [0015]
    [0016] This is due to the fact that the diode batteries have a high parasitic parallel capacitance, the result of all the P-N junctions of the emitting diodes in parallel. Through this parallel parasit capacitor, current flows while it is charging, staying with part of the current that should pass through the laser diode and, therefore, delaying the moment in which the DLS reaches its maximum power. This results in high rise times that limit the achievable pulse time and prevent its use in photoacoustic applications.
    [0017]
    [0018] For example, WO 2014167068 A1 presents a laser diode connected at one end to the drain of a metal-oxide-semiconductor field effect transistor (MOSFET, from the English 'Metal-Oxide-Semiconductor Field-Effect Transistor') and by the another end to a parallel capacitor. However, this circuit operates according to a technique called "quasi-resonant zerocurrent switch." This type of circuit takes advantage of the resonance of the RLC circuit formed by the parallel capacitor, the parasitic resistance and the parasitic inductance of the electric path to give high current pulses in the form of a half-sine This parasitic inductance is mainly due to the tracks of the printed circuit board (PCB), the equivalent series inductance (ESL) The equivalent of the capacitor and the pins of the components are used to generate the pulses, the transistor closes, a resonant oscillation of high current value begins and, when only a half cycle of said oscillation has taken place, the transistor opens, leaving the current that passes through the laser diode. While the transistor is open, the capacitor and parasitic inductance are charged.
    [0019]
    [0020] This configuration has the advantage that the energy efficiency achieved is very high, because it is used to give the pulse of high current, not only the energy stored in the condenser, but also the energy that has been stored in the inductance parasitic of the circuit. However, it has the drawback that the range of pulse times that can be achieved is limited by the resonance of the RLC circuit, presenting less versatility for the described applications.
    [0021]
    [0022] Specifically, there are two requirements that the duration of the optic pulse must fulfill in optoacoustic tomography applications:
    [0023] • Stress confinement requirement: consists in that the pulse must be shorter than the time it takes for the ultrasound wave to propagate through the absorber:
    [0024]
    [0025]
    [0026]
    [0027] Dp being the size of the absorber (equivalent to the maximum resolution of the tomographic image) and vs the speed of sound in the middle.
    [0028] • Thermal confinement requirement: consists of the fact that the pulse duration must be less than the time it takes the absorber to dissipate the calorific energy acquired by the optical pulse. To calculate the maximum time that meets this requirement the following expression is used,
    [0029]
    [0030]
    [0031]
    [0032] where L is the linear length of the absorption (typically, for media where the absorption is greater than scattering, otherwise, the length of the absorber in the direction of absorption) and aT is the thermal diffusivity of the medium. In general, the stress confinement requirement is more restrictive.
    [0033]
    [0034] The frequency of repetition only affects the acquisition time of the system that receives the optoacoustic signal and can be anyone who respects the correct acquisition of the ultrasonic signal and the interference with possible echoes. In addition, the pulse width in emission must be respected. Theoretically, it can be comprised between less than 1 Hz and several MHz. In practice, solid state lasers, conventionally used in optoacoustic, have a frequency of repetition of a few Hz; while high power laser diodes can operate at several kHz. This is the main reason why that the use of diode batteries in optoacoustic is proposed, since it allows, on the one hand, to make the acquisition of data more quickly and, on the other hand, to perform a higher averaging of the obtained samples to obtain a better signal with better signal-noise ratio (SNR).
    [0035]
    [0036] Finally, the power (P) of the optical pulse depends on the energy per pulse (E) that is desired, according to the relationship:
    [0037] P = E -tp
    [0038] The energy (E) required depends on the variation of pressure that can be detected, the Gruneisen parameter (r), the absorption coefficient of the material (^) and the area over which it is irradiated (A):
    [0039]
    [0040]
    [0041]
    [0042] The greater the power, the more penetration will be achieved in the tissues and the optoacoustic signal will be of greater pressure and, therefore, easier to detect.
    [0043]
    [0044] Therefore, there is still a need in the state of the art for an alternative method and control system for light emitting diode batteries capable of generating ultra-high-power pulses that allow the use of said diode batteries in biophotonic applications.
    [0045]
    [0046] Description of the invention
    [0047]
    [0048] The present invention solves the problems described above by combining a N-channel MOSFET transistor and at least one high capacity capacitor that is charged during the periods in which the transistor is closed, avoiding the limitations caused by the resonance frequency of the circuit RLC formed by the parasitic resistance and inductance of the electric path.
    [0049]
    [0050] In a first aspect of the invention there is presented a controller that is connected to a diode stack through a first control port and a second control port, the diode stack having a parasitic resistance (R) and a parasitic inductance ( L). The controller is configured to generate a plurality of pulses with a pulse duration measured at half height (tp). This pulse duration tp meets the thermal confinement requirements and confinement of stress previously mentioned for an opto-active application. To generate said pulse emission, the controller comprises at least the following elements:
    [0051] - A N-channel MOSFET transistor, whose drain is connected to the second control port. Preferably, the controller also comprises a driver connected to the door of the MOSFET, the driver having rise times less than or equal to the rise times of the transistor.
    [0052] - At least one first capacitor connected to the first control port, said capacitor (or the combination in parallel of several first capacitors) having a capacity (C1) greater than or equal to:
    [0053]
    [0054]
    [0055]
    [0056] where, Req is an equivalent resistance of an electric path between the supply voltage (vDD) that includes the parasitic resistance (R) of the diode stack. In particular, said equivalent resistance is preferably:
    [0057] Req = ESR + RdSon + ^ 3
    [0058] where ESR is the equivalent series resistance (ESR, the English 'Equivalent Serial Resistance') of the first capacitor C1 and first capacitors, RD are is the resistance between drain and source of the MOSFET and a resistor R3 is a monitoring port.
    [0059] For this calculation, it is assumed that the first capacitor is charged at the moment immediately before the current pulse, so the load voltage of the first capacitor (VC1) is equal to the supply voltage (vDD). The current that circulates through the laser (ILD), or what is the same, through the electric path previously mentioned is calculated as:
    [0060]
    [0061]
    [0062]
    [0063] where VLD is the drop voltage in the laser diode. The maximum of the laser current is:
    [0064]
    [0065]
    [0066]
    [0067] and the pulse of current will follow the expression:
    [0068]
    [0069]
    [0070] The threshold of the aforementioned first capacitor therefore ensures that the pulse falls to half the height at the desired time.
    [0071] More preferably, the at least one first capacitor is selected so that:
    [0072]
    [0073]
    [0074]
    [0075] where f is the resonance frequency of the RLC circuit formed by the first capacitor, the parasitic inductance (L) and the equivalent resistance (Req), ie:
    [0076]
    [0077]
    [0078]
    [0079]
    [0080] Preferably, the controller further comprises all or a subset of the following resistors:
    [0081] - A first resistance between the driver and the door of the MOSFET. This first resistance protects the MOSFET and has a reduced value, typically recommended by the manufacturer of the MOSFET, such as 1 Q.
    [0082] - A second resistance between the door of the MOSFET and ground. It also protects the MOSFET. If due to some error or component failure, the MOSFET door is left on the air, a capacitive divider is generated between the CGD and CGS loads of said MOSFET and an uncontrolled current through it. This is avoided by placing the second resistance that verifies:
    [0083]
    [0084]
    [0085]
    [0086] where CGS is the load between door and source and CGDes the load between door and drain of the MOSFET.
    [0087] - A third resistor connected to the MOSFET source and to the monitoring port, which defines the proportionality between the current flowing through the laser diode and the monitoring voltage value, and which is part of the equivalent resistance.
    [0088] - A fourth resistor connected to the power supply and to the first capacitor, which acts as a load resistance of said first capacitor, allowing its charge in the time that the MOSFET is open. Therefore, given a repetition frequency frep and a pulse time tp:
    [0089]
    [0090]
    [0091]
    [0092] where T is the period between pulses and tc is the maximum charging time for the capacitor. The theoretical charging time of the capacitor would be infinite given that during charging the voltage of the first capacitor has the expression:
    [0093]
    [0094]
    [0095] However, it can be considered by convention that the capacitor is charged in a time:
    [0096] tc> 5 / 4Ci
    [0097] So the fourth resistance of the controller will verify:
    [0098]
    [0099]
    [0100]
    [0101]
    [0102] Also preferably, the controller is implemented on a printed circuit board with ground plane on at least two sides comprising the tracks of the first control port and the second control port, allowing to minimize the distance to ground of all the components connected to it. said clues. More preferably, the tracks connected to the first control port and the second control port have a greater width than the rest of the tracks, thus reducing their parasite inductance.
    [0103]
    [0104] In a second aspect of the invention there is presented a method of controlling a diode stack comprising generating a pulsed signal with pulse duration (tp) between a first control port (vd1) and a second control port. For said generation, the method comprises at least the following steps:
    [0105] - Switch a MOSFET transistor connected to the second control port through the drain.
    [0106] - When the transistor is open: charge at least a first capacitor connected in parallel to the first control port. Said at least one first capacitor has a capacity (C1):
    [0107]
    [0108]
    [0109]
    [0110] with
    [0111]
    [0112] Req = ESR + RdSon + ^ 3
    [0113] Preferably, the charge of the first capacitor is carried out through a fourth resistance that verifies:
    [0114]
    [0115]
    [0116] Also preferably, the inverse of the resonance frequency (f) of the RLC circuit formed by the at least one first capacitor, the parasitic inductance and the equivalent resistance verifies:
    [0117]
    [0118]
    [0119]
    [0120] - When the transistor is closed: release the charged voltage in the at least one first condenser.
    [0121] - Preferably, monitor the diode stack through a third resistor connected to a transistor source and to a monitoring port.
    [0122]
    [0123] In a third aspect of the invention there is presented a diode stack comprising a plurality of diode bars, as well as a controller according to any of the embodiments of the first aspect of the invention. Note that any preferred option or particular implementation of the controller of the invention can be applied equally to the method and to the stack of diodes of the invention. Also, the elements of said controller can be adapted or configured to implement any step of the method of the invention, according to any particular implementation of both.
    [0124]
    [0125] The controller, control method and diode stack of the invention thus allow to obtain pulses of high power and low pulse time, avoiding the limitations imposed by the resistance and the parasitic inductance of the electric path in conventional techniques. These and other advantages of the invention will be apparent in light of the detailed description thereof.
    [0126]
    [0127] Description of the figures
    [0128]
    [0129] In order to help a better understanding of the characteristics of the invention according to a preferred example of practical realization thereof, and to complement this description, the following figures, whose character is illustrative and are accompanied as an integral part thereof, are accompanied as an integral part thereof. non-limiting:
    [0130]
    [0131] Figure 1 shows a diagram of a diode battery known in the state of the art.
    [0132]
    [0133] Figure 2 shows schematically an operation curve of a diode stack, as well as the comparison between a typical operating point and an operation point of photoacoustic applications.
    [0134]
    [0135] Figure 3 exemplifies the peak power reduction caused by reducing the pulse time with the techniques known in the state of the art.
    [0136]
    [0137] Figure 4 presents schematically the type of target pulse, in which the maximum power is maintained by reducing the pulse time.
    [0138] Figure 5 exemplifies the inputs and outputs of the controller of the invention in accordance with a preferred embodiment thereof.
    [0139]
    [0140] Figure 6 presents the elements that internally conform the controller of the invention according to a preferred embodiment thereof.
    [0141]
    [0142] Figure 7 illustrates the internal operation of the MOSFET transistor driver used by a preferred embodiment of the controller of the invention.
    [0143]
    [0144] Figure 8 schematically shows the connection tracks between elements recorded on the base plate according to a preferred embodiment of the invention.
    [0145]
    [0146] PREFERRED EMBODIMENT OF THE INVENTION
    [0147]
    [0148] In this text, the term "includes" and its derivations (such as "understanding", etc.) should not be understood in an excluding sense, that is, these terms should not be interpreted as excluding the possibility that what is described and defined can include more elements, stages, etc.
    [0149]
    [0150] Figure 2 shows an example of the voltage curve (VLD) - intensity (ILD) of a diode stack, in which a typical point of operation (IT | P-VT | P) is indicated, compared to a point of operation suitable for photoacoustic applications (IOP-VOP), achieved by the present invention. As can be seen, the photoacoustic applications require pulses of current much higher than that obtainable by conventional laser diode control techniques known in the state of the art.
    [0151]
    [0152] Figure 3 shows what happens when the pulse time is reduced to the levels suitable for photoacoustic applications with the known drivers in the state of the art (such as a zero-current quasi-resonant commutator scheme). In particular, an example of a pulse generated by the known drivers in the state of the art is shown with a first pulse time (tP1) and a first pulse power (PP1); compared to a natural pulse of the same diode stack, with a second pulse time (tP2) and a second pulse power (PP2). As can be seen, the initial ascending slope in both cases is the same, but since the rise time (inversely related to the bandwidth) is much greater than the pulse time, the peak power reached is also lower. In the case of the switch quasi resonant zero current, when the transistor opens after a half cycle to get the first pulse time (tP1) is much lower than the second pulse time (tP2), there is also a significant reduction of the first pulse power (PP1) with respect to the second pulse power (PP2).
    [0153]
    [0154] On the contrary, Figure 4 shows schematically the type of optical pulse resulting from changing the working point of the diode stack (100) to a point with a much higher current value, according to the present invention. By providing more current, the bandwidth is increased and the rise time of the pulse is reduced, allowing to reach the desired power in less time. In this way, optical pulses suitable for photoacoustic (ultra-short and high-energy) are achieved. In the comparative example of the figure, it is possible to reduce the duration until the first pulse time (tP1), but maintaining the second pulse power (PP2).
    [0155]
    [0156] Figure 5 shows schematically the inputs and outputs of a preferred embodiment of the controller (200) of the invention, which in turn implements the steps of a preferred embodiment of the method of the invention. The controller (200) comprises two output ports adapted to connect to a stack of diodes (100) and supply the diode current (id) necessary to generate a train of pulses with a repetition frequency (frep) and a pulse time (tp) We will name these ports the first control port (vd1) and the second control port (vd2). Also, the controller (200) comprises an optional monitoring port (vm) which provides a voltage proportional to the supplied diode current (id). As for the inputs, the controller (200) comprises a first power port (vDD), a second power port (vg) and a tripping signal port (vtr, of the English 'trigger').
    [0157]
    [0158] Figure 6 presents in greater detail the internal components of said controller (200), according to a preferred embodiment thereof. The controller (200) comprises an N channel MOSFET transistor (400), whose drain (D) is connected to the second control port (vd2). Connected to the first control port (vd1), the controller (200) comprises a plurality of first capacitors (C1) connected in parallel. The set of the first capacitors (C1) makes it possible to reduce the resonance frequency of the RLC circuit formed by the first capacitors (C1), the parasitic resistance (R) of the diode battery (100) and the parasitic inductance (L) of said capacitor. diode stack (100). Thus, in the pulse time (tp) in which the diode stack (100) is emitting, the voltage in the capacitor has a quasi-flat response. Also, during the time in which the stack of diodes (100) does not emit, the very first capacitors (C1) accumulate a large amount of charge that they subsequently deliver in high current pulses. Therefore, the first capacitors (C1) are used as an auxiliary battery to the voltage source of the control circuit, being charged to the same voltage as said circuit. By not depending on the pulse time (tp) of the resonance of the RLC circuit, but of the time in which the transistor (400) remains closed, a greater versatility in terms of the pulse regime is achieved.
    [0159]
    [0160] The transistor (400) is controlled in turn by a driver (300). The driver (300) comprises an input port (vi) through which the trip signal that arrives via (vtr), a ground port (vgnd) and a driver power port (vcc) is input. Also, it comprises an output port (vo) that provides the control signal that is input to the gate (G) of the transistor (400). The values provided by the output port (vo) oscillate between two limits that are also introduced in the driver (300) through both ports of minimum value (vmin) and maximum value port (vmax). Note that in the particular implementation shown, the minimum value port (vmin) is directly connected to the ground port (vgnd), while the maximum value port (vmax) is connected to the driver power port (vcc). Also, the connection between the driver power port (vcc) and the second power port (vg) optionally comprises a second capacitor (C2) and a third capacitor (C3) of decoupling in parallel, recommended use in integrated circuits .
    [0161]
    [0162] In order to protect the transistor (400), the gate (G) of the transistor is connected to the output port (vo) of the driver through a first resistor (R1) and to earth through a second resistor (R2) several orders of magnitude greater than said first resistance (R1). Also, the source (S) of the transistor is connected to the monitoring port (vm) by a third resistor (R3). For its part, the first power port (vDD) is connected to the first control port (vd1) through a fourth resistor (R4). At the connection between the fourth resistor (R4) and the first power port (vDD) a second capacitor (C2) and a third capacitor (C3) are also included in parallel.
    [0163]
    [0164] According to an example of particular realization, considering tp = 150 ns, ESR = 33 mQ and RDSON = 10 mQ, the condensers, resistors and feeds described can be implemented with the following values:
    [0165] - C2 = 4.7 | jF.
    [0166] - C3 = 100 nF.
    [0167] - Ri = 1 n.
    [0168] - R2 = i kn.
    [0169] - R3 = 10 mn.
    [0170] - R4 = 10 n.
    [0171] - Vg = 10 V.
    [0172] - V dd = 30 V.
    [0173] - Req = ESR R dson + R3 = 53 mn
    [0174] - C1 = 15 | jF> tp / Req = 2.83 | jF.
    [0175]
    [0176] Note that the plurality of first capacitors (C1) have a low equivalent series resistance (ESR, from the English 'Equivalent Serial Resistance'). The ESR is a value that depends both on the technology and the morphology with which the capacitor is manufactured, and on the frequency and value of the capacity of the same. Said ESR can be calculated as ESR = DF / wC, where w is the frequency, C is the capacity, and DF is a Dissipation Factor defined as the inverse of the quality factor of the resonant circuit, being usually provided by the manufacturer.
    [0177]
    [0178] As for the driver (300), its characteristics are subject to the choice of the transistor (400). Once said transistor (400) has been chosen, the characteristics of this MOSFET will determine the value of the door voltage (VG), roughly between 8 and 12 V. In addition, the parallel of the door-drain capacity (CGD) and the capacity Gate-source (CGS) will determine the peak current needed to charge the gate on each pulse. Typically, a current peak greater than 1 A will be necessary. Therefore, a driver (300) with lower or similar rise times than the transistor (400), which can operate with pulses between 0 V and VG, is selected, and able to give the necessary current to load the door (roughly greater than 2 A).
    [0179]
    [0180] Figure 7 shows the block diagram of a possible implementation of the driver (300) of the transistor (400), commercial drivers known in the state of the art can be used for it. The serial input port (vi) is amplified by an amplifier with hysteresis (370), whose output feeds a first AND gate (380). Said AND gate (380) has a second denied input, to which is connected an output of a first sub-tension blocking module (320, from the English 'undervoltage lockout'). The output of the first AND gate (380) is connected to the input of a level displacement module (390, from the English 'level shifter'). The output of said level displacement module (390) is in turn connected to the input of a second AND gate (340), at whose input The output of a second undervoltage blocking module (330) is connected. Finally, the output of the second AND gate (340) is amplified in an amplifier (350) to obtain the signal that is transmitted to the output port (vo). The driver (300) also comprises a capacitor (360) between the port of minimum value (vmin) and maximum value port (vmax). Also, it comprises a zener diode (310) between the driver power port (vcc) and the maximum value port (vmax).
    [0181]
    [0182] Finally, figure 8 shows schematically the connecting tracks of the elements when installed in a PCB (500). Preferably, it is a class V PCB (500) with ground plane on both sides, BNC connectors for the monitoring port (vm) and the trigger port (vtr), and side extensions (510) for connection to the ports of the diode stack (100). In particular, in an implementation example, it can be implemented in a PCB (500) with an FR-4 dielectric, although the expert can understand that it can also be implemented in other materials, preferably of equal or greater electrical permeability.
    [0183]
    [0184] In order to reduce the inductance of the tracks through which the diode current (id) circulates, the tracks connected to said lateral extensions (510) have a greater width and a minimum length within what is allowed by the morphology of the components . In particular, in the scheme of figure 8, a first track (520) can be observed with a first width (d1) connecting the first control port (vd1) to the first capacitors (C1); a second track (530) with the same first width (d1) joining the first control port (vd1) to the transistor (400); a third track (540) with a second width (d2) joining the fourth resistance (R4) to the first feed port (vDD); and a fourth track (550) with a third width (d3) connecting the second power port (vg) and the driver (300). As can be seen, the first width (d1) is greater than the second width (d2), the second width being in turn greater than the third width (d3).
    [0185]
    [0186] Also, the tracks through which the diode current flows (id) are designed so that they have the ground reference as close as possible to all their points. For this reason, the ground plane is preferably selected on both sides, with a gap between faces as narrow as possible. Alternatively, a substrate of lower height can be used or additional intermediate faces can be inserted.
    [0187]
    [0188] The person skilled in the art will be able to understand that the invention has been described according to some preferred embodiments thereof, but that multiple variations may be introduced in said preferred embodiments, without leaving the object of the invention as it has been claimed.
    权利要求:
    Claims (14)
    [1]
    1. - Controller (200) of a diode stack (100) comprising a first control port (vd1) and a second control port (vd2) adapted to connect to said stack of diodes (100) and generate a plurality of pulses with a pulse duration (tp), the diode stack having a parasitic inductance (L) and a parasitic resistance (R), characterized in that it also comprises:
    - at least a first capacitor (C1) connected in parallel to the first control port ( v d1 ); the capacitance of the first capacitor (C1) being greater than or equal to the pulse duration (tp) divided by an equivalent resistance (Req) of an electric path between power ( v DD) and earth that includes the parasitic resistance (R); Y
    - a transistor (400) of metal-oxide-semiconductor field effect of channel N, with a drain (D) connected to the second control port (vd2).
    [2]
    2. - Controller (200) according to any of the preceding claims characterized in that the inverse of the resonance frequency of the RLC circuit formed by the at least a first capacitor (C1), the parasitic inductance (L) and the equivalent resistance (Req) is greater than at least ten times the pulse time (tp).
    [3]
    3. - Controller (200) according to any of the preceding claims characterized in that it further comprises a driver (300) connected to a gate (G) of the transistor (400), the driver (300) having rise times equal or less than the rise times of the transistor (400).
    [4]
    4. - Controller (200) according to claim 3 characterized in that it also comprises a first resistance (R1) between the driver (300) and the door (G), and a second resistance (R2) between the door (G) and earth, the second resistance (R2) having a value at least two orders of magnitude greater than the first resistance (R1).
    [5]
    5. - Controller (200) according to any of the preceding claims characterized in that it further comprises a third resistance (R3) connected by a same end to a source (S) of the transistor (400) and a monitoring port (vm) .
    [6]
    6. - Controller (200) according to any of the preceding claims characterized in that it also comprises a fourth resistance (R4) connected to the feed ( v DD) and to the at least one first capacitor (C 1 ), the product of the fourth resistance (R 4 ) and the at least one first capacitor (C 1 ) being less than or equal to one fifth of the difference between the time between pulses (T) and the pulse duration (t p ).
    [7]
    7. - Controller (200) according to any of the preceding claims characterized in that it comprises a printed circuit board (500) with ground plane in at least two layers.
    [8]
    8. - Controller (200) according to claim 7, characterized in that it comprises a printed circuit board (500) with a first track (520) between the at least one capacitor (C 1 ) and the first control port ( v d1) ), and a second track (530) between the drain (D) and the second control port ( v D2 ), the first track (520) and the second track (530) having a width (d 1 ) greater than the rest of tracks on the printed circuit board (500).
    [9]
    9. - Controller (200) according to any of the preceding claims characterized in that it is configured to increase the bandwidth of the optical pulses emitted by the diode stack, operating at a working point of greater current than usual.
    [10]
    10. Control method of a diode battery (100) comprising generating a pulsed control signal between a first control port ( v d1 ) and a second control port ( v D2 ) with a pulse duration (t p ), the diode stack having a parasitic inductance (L) and a parasitic resistance (R), characterized in that it comprises:
    - switching a N-channel metal-oxide-semiconductor field effect transistor (400) with a dreamer (D) connected to the second control port ( v D2 ); - when the transistor (400) is open, charge at least a first capacitor (C 1 ) connected in parallel to the first control port ( v d 1 ); the capacity of the first capacitor (C 1 ) greater than or equal to the pulse duration (t p ) divided by an equivalent resistance (R eq ) of an electric path between power ( vdd ) and earth that includes the parasitic resistance (R) ; Y
    - when the transistor (400) is closed, release charge stored in the at least one first capacitor (C 1 ).
    [11]
    11. - Method according to claim 9, characterized in that it comprises further monitor the stack of diodes (100) through a third resistor (R3) connected at the same end to a source (S) of the transistor (400) and to a monitoring port (vm).
    [12]
    12. Method according to any of claims 9 and 10 characterized in that the step of charging the at least one first capacitor (C1) further comprises charging said at least one first capacitor (C1) through a fourth resistor (R4) ), the product of the fourth resistance (R4) and the at least one first capacitor (Cl) being less than or equal to one fifth of the difference between the time between pulses (T) and the pulse duration (tp) .
    [13]
    13. - Method according to any of claims 9 to 11 characterized in that the inverse of the resonance frequency of the RLC circuit formed by the at least a first capacitor (C1), parasitic inductance (L) and equivalent resistance ( Req) is greater than at least ten times the pulse time (tp).
    [14]
    14. - Battery of diodes (100) comprising a plurality of diodes (110) grouped in bars (120), characterized in that it also comprises a controller according to any of claims 1 to 8.
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    同族专利:
    公开号 | 公开日
    WO2019077187A1|2019-04-25|
    ES2710080B2|2019-09-18|
    引用文献:
    公开号 | 申请日 | 公开日 | 申请人 | 专利标题
    US20140063593A1|2012-08-31|2014-03-06|Martin Ole Berendt|Capacitor discharge pulse drive circuit with fast recovery|
    WO2014150730A1|2013-03-15|2014-09-25|Raytheon Company|Diode driver for battery-operated laser systems|
    法律状态:
    2019-04-22| BA2A| Patent application published|Ref document number: 2710080 Country of ref document: ES Kind code of ref document: A1 Effective date: 20190422 |
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    优先权:
    申请号 | 申请日 | 专利标题
    ES201731228A|ES2710080B2|2017-10-18|2017-10-18|Controller and control method of a diode stack|ES201731228A| ES2710080B2|2017-10-18|2017-10-18|Controller and control method of a diode stack|
    PCT/ES2018/070676| WO2019077187A1|2017-10-18|2018-10-17|Controller and method for controlling a diode stack|
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