瑞士专利CH713962B1 Method and device for determining phase errors or clock delays in a measuring transducer and power m

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专利摘要:
Current transformers are widely used in current measurement systems. They provide good insulation between the supply voltage and the measuring device. However, they can introduce small phase errors which can become significant sources of error if the current to a load (24) is out of phase with the supply voltage for the load (24). The present invention relates to a robust measurement apparatus and method that can be used in situ to monitor for and correct phase errors. The invention also relates to a power meter.
公开号:CH713962B1
申请号:CH01549/18
申请日:2016-12-21
公开日:2021-10-29
发明作者:Ephraim David Hurwitz Johnatan；Amir Ali Danesh Seyed；Michael James Holland William；Ye Shaoli
申请人:Analog Devices Global；
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PRIORITY CLAIM
The present application claims the priority benefit of U.S. Patent Application No. 15 / 182,094, filed June 14, 2016, which is hereby incorporated by reference in its entirety.
AREA
The present invention relates to methods and apparatus for estimating a phase shift in measurements of a measured quantity occurring in a measurement transducer, such as a current transformer, and relates to power measurement systems having such an apparatus. Such a phase shift from the transducer can be viewed as a phase measurement error. The teachings of the present invention can also be used to evaluate delays and phase shifts resulting from signal processing chains.
BACKGROUND
There is often a desire to know the electricity that is being supplied to "a user" of the electricity, which user could be a factory, a distribution circuit within a factory or a dwelling, or one or more facilities. It is also often very desirable to know the actual amount of energy being used by "the user" so that a utility company can bill the user for the amount of energy they are using.
The power consumed by a device that is supplied with a sinusoidal voltage and draws a sinusoidal current can be calculated from the following:P = V * I * Cosθ Equation 1where: V is the voltage, I is the load current, and θ is the phase angle between the voltage waveform applied to the device and the current flowing in the device. Cosθ is known as the power factor.
As is known to one skilled in the art, the angle θ represents the phase difference between the applied voltage waveform and the resulting current due to inductive or capacitive loads fed by the power line. In a simple case it is assumed that both are sinusoidal at the line frequency. In such a simple system it is then relatively easy to operate in terms of phase shifts. In reality, however, a load, such as a motor, switched mode power supply, or inverter, may have a complex current draw that has components at multiples of the line frequency and / or a switching frequency in the load.
Furthermore, regulators often require that consumers be treated fairly by their energy suppliers and thus strict tolerance limits are imposed on the accuracy of power meters (watt-hour meters). It is therefore important that such measuring devices maintain high levels of accuracy under all expected operating conditions. This means that the line voltage must be measured, the line current must be measured, and that any phase difference between the line voltage and line current measurements must be accounted for with sufficient accuracy that the estimate of the power consumed by a load is within the specified degrees of accuracy falls.
The converters can introduce errors. For example, the current converter, such as a current transformer, may introduce an error in the magnitude of the current being measured. It can also introduce a phase error in the estimate of the phase between the voltage and the current, both of which can be represented on a phasor diagram. These errors can negatively affect the accuracy of estimates of power consumed by a load. Similarly, filters, for example filters used to discard glitches, can introduce a delay in the current and voltage processing signal paths. Furthermore, while the average delay through the filters for a set of products, such as power meters, may be known to an acceptable accuracy on a statistical basis, component-to-component manufacturing variation may mean the absolute delay, or indeed the Frequency response of any given filter is not known.
It is desirable to be able to estimate the phase error introduced by a current transformer and / or other signal processing components within a signal processing chain involved in measuring the voltage and current within a watt-hour meter.
SHORT REPRESENTATION
According to the first aspect of the present invention, a method for estimating phase measurement errors in measurements of a measured quantity, such as current or voltage, is provided. The method comprises providing an input signal with a fundamental frequency to an input of a measuring transducer, the measuring transducer serving to measure measured variables. For example, if it is desired to measure current, then the input signal is applied to a current transformer. The input signal is not sinusoidal.
The input signal can be a repetitive signal. The repetitive signal preferably, but not necessarily, has nominally linear rising and falling edges. This makes it relatively easy to create. The repetitive signal can be generated, for example, by an inexpensive square wave generator or by a relatively inexpensive low-precision digital-to-analog converter.
An output signal from the transducer is received. The output signal from the transducer is analyzed to determine a phase difference compared to the input signal. The analysis may include correlating the input and output signals to determine a propagation delay and, therefrom, a phase error.
The method further comprises applying a first correction to the determined phase difference in order to compensate for a finite rate of change of the input signal. This determines an estimated phase measurement error in the measurement of the measured variable.
The phase measurement error can then be used by a power meter to improve the accuracy of a power estimate.
It is highly desirable that the method of measuring phase errors be relatively simple and not cumbersome in terms of computation or measuring equipment required, so that an apparatus implementing the method can be included in a device such as a watt-hour meter can without incurring excessive costs. Preferably, the measurement of the error is repeated so that the device reacts to changes in the phase measurement error introduced by the current transformer, for example as a result of a change in temperature, the resistance of windings of the current transformer or the magnetic properties of the core, such as changes in the Permeability (which is a complex variable) of the core is influenced by the frequency and size of the magnetic field generated by the measured current and / or from stray magnetic fields. The measurement of the error can be carried out repeatedly or continuously in accordance with a measurement schedule.
A significant potential cost burden is the apparatus required to generate the input signal. Signal generators capable of generating high quality sinusoidal signals tend to be relatively expensive devices. For a mass produced product operating in a competitive cost environment, the use of such expensive signal generators is ultimately excluded for economic reasons. It is therefore desirable to find a way to use lower cost signal generators without sacrificing measurement performance.
In accordance with the teachings of the present disclosure, a high quality sinusoidal input signal is not required. The input signal can be generated by digital electronics or can have a digital-like form. In its simplest form, the input signal can transition between first and second levels in a predictable pattern. Such a signal can be a square wave or at least a square wave. The signal does not have to have a 50-50 mark-to-space ratio. Square wave generators are much cheaper to implement than other forms of signal generator. However, even for a square wave generator, there are practical considerations that significantly affect its implementation cost. An ideal square wave immediately transitions between a high voltage or high current state and a low voltage or low current state, or generally between a first and second state. Real-world drive circuits, however, have finite rates of voltage change or current change because they are either bandwidth-limited or slew rate-limited in operation. The square wave generator may also suffer from overshoot or undershoot and may have an asymmetrical output waveform. The inventors have recognized that a method for determining a phase measurement error would have to take into account transitions that are limited in rate of increase or bandwidth or other imperfection artifacts in the control signal. Once the error is determined, steps can then be taken to correct it or otherwise mitigate its effects.
This implementation enables non-ideal versions of square wave-like signals to be tolerated or even intentionally adopted for ease of characterization and implementation. For example, signals can be selected in which transitions are defined by exponential functions. Such functions occur when a capacitor is charged or discharged through a resistor and are inexpensive to implement and have a reliable waveform due to the simplicity of the components used.
The input waveform can be generated by a digital-to-analog converter or DAC. This enables discrete / step approximations of waveforms of any desired shape to be generated. Such waveforms can approximate sinusoids, triangular waves, square waves, or irregular shapes. The shapes can be modified to reduce the load on drive amplifiers or to modify the frequency spectrum of the drive signal to reduce the risk of interference. Alternatively, the DAC can be driven by a random or pseudo-random input sequence so that the drive signal looks like noise. However, autocorrelation techniques can be used to identify the input signal and estimate the time delay it will experience as it propagates through the signal processing chain. The input signal can be subjected to filtering in the analog or digital domain if a delay is to be analyzed and characterized as a function of frequency.
In an extension of this technique, a less defined source can be used to generate the input signal, or a naturally occurring signal on a live conductor can be used in place of the signal generator and can be digitized by an analog-to-digital converter, which can bypass the anti-aliasing filter to capture the noise signals and then correlate the output signal with the input signal to estimate the propagation delay and / or phase error introduced by a transducer or signal processing path. Alternatively, if the glitch filters are common to all signal paths (and are well matched, for example, being formed on the same silicon die in an integrated circuit), then the absolute delay introduced by the filter need not be known as it cancels out in the power calculations, in which case the ADC can work with a filtered signal.
The present disclosure is not limited to the use of two-level digital signals. Other non-sinusoidal signals can be used.
A second aspect of the present invention relates to an apparatus for estimating phase measurement errors in measurements of measured quantities. The device has a signal generator for providing an input signal to an input of a measuring transducer, which is used to measure a measured variable, such as a repetitive input signal with a fundamental frequency. Where the input signal is used to modulate a measured quantity, such as a current measured by a transducer such as a current transformer, and wherein the repetitive signal is not sinusoidal or at least not a high quality sinusoid. The apparatus further includes a phase and time shift comparator for receiving an output signal from the transducer and for analyzing the output signal to determine a phase or time difference compared to the input signal. The phase and time shift comparator is designed to determine an estimated phase shift by applying a phase correction to the determined phase difference in order to compensate for imperfection artifacts in the input signal due to a finite rate of change of the input signal.
The apparatus may be included in a power meter that is used to estimate the amount of energy consumed or transferred to an electricity distribution system.
The apparatus and method can also be used during the manufacture of a transducer or watt-hour meter in order to characterize the transducer or to calibrate the measuring device. In the case of a measuring device, the calibration values can be stored in a memory within the measuring device. The meter may further include communication means (which may be wired or wireless) to send data such as the power that has been consumed and the estimate of a phase measurement error to a remote party such as a utility. This enables a health check of the measuring device to be carried out, since drift or degradation can be monitored. It can also provide information regarding attempts to tamper with the current measurement circuitry.
BRIEF DESCRIPTION OF THE DRAWINGS
Embodiments of the present invention will now be described by way of non-limiting example only with reference to the accompanying figures, in which: Figure 1 illustrates schematically the data acquisition channels of an electronic watt-hour meter; Figure 2 schematically illustrates a current transformer; Figure 3 is the circuit diagram representation of the device shown in Figure 2; Figure 4 is a graph showing the percent error in power measurement for 1 degree and 2 degree errors in estimating current-to-voltage phase shift θ for various power factors; Figure 5 is an equivalent circuit for the current transformer of Figure 2; Figure 6 is a simplified equivalent circuit; Figure 7 schematically illustrates a first embodiment of the present invention; Figure 8 schematically illustrates a second embodiment of the present invention; Figure 9 schematically illustrates the waveform of a drive signal used in embodiments of the present invention; Figure 10 illustrates one embodiment of a circuit for detecting the time for the square wave to transition from high to low currents; Figures 11a through 11c illustrate waveforms in the circuit of Figure 10; Figure 12 is a circuit diagram of a current modulator that can be used in an embodiment of the present invention; Figure 13 is a circuit diagram of a watt hour meter in accordance with the teachings of the present invention; FIG. 14 is a flow diagram of a method for estimating a phase measurement error; FIG. 15 is a flow diagram of a method for correcting a phase measurement error; Figures 16a through 16d illustrate how the ideal square waveform can be distorted; Figure 17 is a schematic diagram of a measurement system for indicating how delays can accumulate; and Figure 18 is a schematic diagram of another embodiment of the present invention.
DESCRIPTION OF SOME EMBODIMENTS OF THE INVENTION
It is often desirable to measure electrical parameters such as a voltage supplied to a load and / or a current supplied to a load. In order to provide a more accurate assessment of the drawn power based on an actual voltage and an actual current as opposed to assuming a nominal voltage and a sinusoidal load current, it is known to use digital measuring devices.
Figure 1 shows in schematic form the main components in the data acquisition channels for a digital power meter. The power from a power supply, such as a mains supply, is provided to a load. The load can be a single phase load or a multi-phase load. For the or each phase, the load voltage is measured by a suitable voltage sensor 2 and the line current for the or each phase is measured by a suitable current sensor 3.
The output from the voltage sensor 2 (or from the sensors in a polyphase system) can be passed through a filter 4 to limit the signal bandwidth to a suitable range, for example to avoid aliasing, and then to an analog-digital Converter 5 are given. The analog-to-digital converter (ADC) may be associated with a programmable gain amplifier. Likewise, the output from the or each current sensor 3 may be passed through a suitable filter 6 and then digitized by an ADC 7 which may be associated with a programmable gain amplifier.
The voltage sensor is often a potential divider, so the response of the potential divider should be quick, i. H. it introduces no processing delay or phase error. The anti-aliasing filter will introduce a phase delay and although this should be well determined, tolerances during manufacture mean that its delay is likely not to be known precisely as its cutoff frequency will only be known roughly. The ADC will also introduce a delay, but since this is a digital component, its performance is determined by system clocks.
Similar comments apply to the current measuring channel as to the above for the filter 6 and the ADC 7. However, the current measuring sensor can introduce a further phase error / a further delay depending on the current measuring technology used. Shunt resistors do not introduce a phase delay, but have the disadvantage that they have to be placed in the supply path. On the other hand, current transformers can be placed in situ around a conductor and have excellent insulation properties. However, they introduce phase delays.
To put the present disclosure in context, it is useful to consider the operation of a current transformer. Figure 2 schematically illustrates the components of a current transformer. Basically, a conductor 10 carrying an alternating current to be measured acts as the primary winding of the current transformer. A secondary winding 12 is magnetically coupled to the primary winding 10. The secondary winding 12 can be wound around the primary winding 10 or can be wound around a core 14 that is magnetically coupled to the primary winding 10. Due to their nature, current transformers provide good insulation between conductor 10 and secondary winding 12. They also have minimal impact on conductor 10 and if core 14 can be split the current transformer can be inserted around primary conductor 10 without it to separate.
The effective turns ratio between the primary winding and the secondary winding is normally given by the ratio of the current that flows through the primary winding to the current that is output through the secondary winding. A 1000 to 1 ratio transformer would output 1 amp from the secondary for every 1000 amps flowing through the primary. The transformers can be tapped to allow the measurement circuit following the transformer to operate over a wider range of currents. A physical device shown in FIG. 2 can be represented by the circuit diagram of FIG.
As previously noted, the consumer must be accurately billed for the amount of power he uses. Too high a computation is unacceptable to regulators, and too low a computation represents a potential loss of revenue. A significant problem is that even small phase errors can result in large errors in measuring the amount of power consumed.
As previously noted, the power dissipated is a function not only of voltage and current, but also of the phase θ between the voltage and the current.
It is known that due to the inductance and resistance in the current transformer, the current transformer itself introduces a phase error. Thus, the measured power can be represented as Pmess, wherePmess = V * (I * Ki) * Cos (θ + α). Equation 2
Whereas the actual performance is:Actually = V * (I * Ki) * Cosθ. Equation 3
Where Ki represents a scaling factor for the current transformer and α represents a phase error introduced by the current transformer.
If we only focus on errors introduced by the phase error, the error can be represented by:Error = (Ptactual- Pmess) / Ptactual = 1- (Cos (θ + α) / Cosθ) Equation 4
As a result, when the power factor is high (close to unity) the influence of the phase error on the measurement is small or negligible. However, as the power factor increases, the influence of the phase error increases significantly.
A graph which shows the power measurement error as a function of the power factor for the one-degree phase error and a two-degree phase error is shown in FIG. It can be seen that for power factors of 1 (the voltage and current are in phase) a 2 degree error in the phase measurement is therefore not a problem. However, if a load has a power factor of 0.6 (θ = 53 degrees), then a 2 degree error in the phase measurement manifests itself as a 5% error in the power measurement. It is therefore desirable to accurately characterize the phase error of a current transformer and, in fact, the signal processing chain associated with the current transformer.
A problem with current transformers is that their answer is potentially quite complex. Figure 5 is an equivalent circuit diagram of a power converter. The primary winding resistance is denoted by Rp, the primary winding inductance is Lp, the secondary winding resistance is Rs, the secondary winding inductance is Ls and the load resistance is Zb. Zm is the magnetization impedance of the transformer (for example, the transformer may have a magnetic core). We can generally ignore Rp and Lp and express the variables occurring on the primary side of the transformer in terms of equivalent quantities as the secondary winding, then the circuit can be represented as shown in FIG. I '= Ip / α, where α is the transformer ratio; Z'm = Zm / α <2> and Im '= Im / α. The angle between I 'and Ib is the phase error introduced by the current transformer. We can note that I '= I'm + Ib; E2 = I'mZ'm; and E2 = Ib (Rs + jωLs) + Ib (rb + jωxb), where ω is the angular frequency in radians per second. This can be used to provide insight into the fact that the phase error changes with the magnitude of the load resistance. However, it also tells us that the relative sizes of the real and imaginary parts of impedance also change with frequency. While the line frequency is generally at a known frequency, e.g. B. 50 Hz, 60 Hz, 400 Hz (aircraft), loads such as inverters can be a source of higher order harmonics that should be considered if an accurate assessment of the delivered power is to be achieved. The magnetization impedance can change with both frequency and load current.
It is advantageous to be able to check the response of the current transformer. This could be done by providing a very pure sinusoidal signal as a perturbation of the current through the current transformer and then performing a frequency extraction of this signal (generally using Fourier analysis). This requires that the cost and effort at the signal source and computational effort be made to perform an FFT analysis. It would be advantageous to use less expensive signal sources such as slew rate limited square wave generators. These are easy to manufacture, for example using a digital inverter in a ring, or by switching a logic gate in response to a counter / timer or a signal from a data processor implementing a numerically controlled square wave oscillator as one of its tasks. The signal does not need to have a 50-50 mark-to-space ratio and this can further simplify the circuits that generate it. Likewise, the rate of increase in a voltage or current increase direction (pull-up direction) need not coincide with the increase rate in a voltage or current decrease direction (pull-down direction). Other performance limitations are discussed later.
FIG. 7 schematically illustrates a current measuring device according to a first embodiment of the present disclosure. A conductor 20 enables current to flow between a first node 22 and a second node 24. The first node 22 can be connected to an AC power supply and the second node 24 can be connected to a load. However, under certain circumstances, load 24 can both consume and deliver energy. For example, node 24 could represent a residential building that generally consumes energy, but would also have photovoltaic panels, so that the residential building can supply energy back to the electrical supply network represented by node 22 when the photovoltaic panels generate more energy in use than the residential building requires . The current flowing through the conductor 20 can be measured by a current transformer 30. The current measuring circuit of FIG. 7 can also be connected to a voltage measuring circuit 32, so that the actual power which is supplied from the node 22 to the node 24 can be measured by a measuring circuit 50, for example for billing purposes.
Watt hour meters used for electrical metering for billing purposes are typically specified to be 0.5% or 1% accurate. It can therefore be seen that even a small phase error of less than 1 degree is unacceptable even for power factors of around 0.9. Residences can have a power factor other than unity due to the use of fluorescent lighting, washing machines, induction ovens, and so on. Industrial facilities are more likely to have large inductive loads, but they are equally likely to have power factor correctors installed to cut their energy bills.
Nonetheless, it can be seen that it is highly desirable, in order to meet the accuracy standards required for watt hour meters, to compensate for any phase errors in the current sensing transformer 30. In the arrangement shown in FIG. 7, a current modulation circuit 60 is provided so that the current drawn from node 22 is modulated in a known manner. The current modulation circuit 60 can be connected directly to the conductor 20 and can periodically switch between drawing a first known current level and a second known current level, one of which levels can be zero current flow. This switching information is provided to the measuring circuit 50, which responds to the output of the current transformer 30 and which can compare the times at which the modulated current 40 changes with observations of this change, as made by the current transformer 30, in order to determine a phase error of the current transformer 30 estimate. The timing of the changes in the modulated current can be controlled by a controller 62, which can also provide timing data to the measurement circuit 50. To simplify the circuit, estimates of the phase measurement error can be limited to be performed in one of the voltage half-waves of the mains supply.
Figure 8 illustrates an alternative arrangement to that shown in Figure 7, in which the current modulation circuit 60 is not directly connected to the conductor 20, but instead controls the modulated current through a further conductor 64, which is adjacent to the conductor through the current transformer 30 20 is running. This arrangement ensures galvanic isolation between the conductor 20 and the driver circuit 60 for the modulated current. Otherwise, the operation of the circuit is similar to that of the circuit described with reference to FIG.
In order to enable an implementation of an inexpensive and reliable drive circuit 60 for the modulated current, the drive circuit 60 supplies a square wave current for the modulated current. The square wave current is illustrated schematically in FIG. The square wave current can be achieved, for example, by selectively turning a current source on and off, or by putting a current source in association with a current steering circuit, as will be described later. Whichever approach is taken, however, the parasitic capacitive and inductive components associated with drive circuit 60 and conductor 20 or conductor 64 will be such that the current does not transition immediately between a first value 70 and a second value 72 . If the drive circuit for the modulated current switches between the first value 70 and the second value 72, then although the switch design may result in the current change being initiated at time T1, the current will instead be increased due to its being rate limited do not reach the second value 72 until the time T2. The time difference T2 - T1 has significant effects on a subsequent estimate of the phase measurement error from the transformer. Likewise, if the square wave transitions from the second value 72 to the first value 70, although the transition may start at time T3, it will not end until time T4. Additionally, if the drive circuit 70 includes active circuitry, such as an amplifier, the active circuit / amplifier may have a finite gain bandwidth or slew rate limit that significantly affects the circuit response. The response of the drive circuit 70 may therefore vary as a function of temperature, or may vary over its life, or in fact may vary with manufacturing variations. Therefore, the actual shape of the drive signal may not be accurately known.
The inventors have recognized that any phase measurements that are estimated as a result of the application of the nominal square wave drive signal to the current in conductor 20 or its flow in measurement conductor 62 must take into account the time it takes for the square wave between the first value 70 and second value 72 transitions, and any estimates of phase change must be made with reference to an appropriate value such as the midpoint of the transition, i. H. 1⁄2 (T1 + T2) and 1⁄2 (T3 + T4), and not the nominal start times T1 and T3. Furthermore, applying this correction means that the power required by the square wave generator is not as critical, so smaller and less power hungry devices can be used.
The duration of the transition, for example from T1 to T2, can be estimated by starting a counter at T1 and stopping the counter when it is determined that the second current value 72 has been reached at time T2. The value of the count held in the counter can then be converted into a time offset and supplied to the measuring circuit 50 as a corrected transition signal.
The correction to limit the rise rate can be carried out using an estimation circuit 80, as shown in FIG. For example, the estimation circuit 80 may include a relatively low value resistor 90 inserted in the current flow path to and from the current modulator circuit 60 so that the modulated current is measured. The voltage developed across resistor 90 can be DC-blocked and amplified by an amplifier 92, which has a moderate high-pass filter response, and then filtered by a further high-pass filter 94. The output of filter 94 selects the edges of the square wave. The voltage can then be detected by rising and falling edge detectors 96 and 98, respectively, to start and stop a counter timing so that the time for the square wave to transition between its first and second current values is accurately measured and thus is measured the midpoint of the square wave transition is accurately estimated and this information is provided to the measurement circuit 50 so that it can correctly take into account the phase angle between the voltage and current delivered from node 22 to node 24 to determine the amount of energy correctly identify that is being used by a device connected to node 24.
The signals in the circuit of FIG. 10 are shown in greater detail in FIGS. 11a to 11c. Figure 11a illustrates the voltage developed across resistor 90 as the current from the current modulator changes from a relatively high value to a low one. The voltage across the resistor may be high pass filtered to remove the DC component amplified by amplifier 92 to obtain a pulse-like shape as shown in Figure 11b. This is then passed through the high-pass filter 94 in order to identify the edges 100 and 102, as shown in FIG. 11c, which can be detected by the edge detectors 96 and 98 in order to start and stop a clock. Although this functionality has been described in the analog domain, the same result can be achieved by digitizing the voltage across the resistor and analyzing the digital sample to look for the edges of the current transitions.
The current flow could be bipolar (i.e., both positive and negative) or it could only be unipolar. Unipolar is easier to achieve because this can be done using a current mirror, as shown in FIG. The current mirror 120 is known to those skilled in the art and the current flowing in transistor 122 is copied through transistor 124, according to a scaling factor for modulating the current in conductor 20 or 64. The current in transistor 122 can be established by taking the voltage output of a Counter is taken and converted into a current by passing through resistor 132. The counter is expediently a divide-by-2 counter, since this acts to clean an incoming clock into a square wave with a uniform mark-to-space ratio. As an alternative to this, the current mirror could be activated by a digital-to-analog converter under the control of the controller 62.
After the spurious current is established and the midpoint of each transition identified, these midpoints can be compared to corresponding changes in the current measured by the current transformer to determine how much phase error the current transformer introduces.
FIG. 13 illustrates a further embodiment of a power meter 150 which is associated with a first supply conductor 152 which extends between a first supply node S1 and a first load node L1. A second supply conductor 152 extends between a second supply node S2 and a second load node L2. The second conductor, which can be a live conductor, is a single phase supply, but the teachings herein are extendable to, for example, three phase supplies.
A current transformer 160 has a coil that is coupled to the second supply conductor 164 and also to an excitation current generated by a phase error measurement circuit 170. The current at the output of the current transformer is converted into a voltage by a load resistor 172 and the voltage across the resistor 122 is digitized by an analog-to-digital converter 174. The output of the analog-to-digital converter 174 is a stream of samples Is, where S is an index and S varies as a function of time.
A potential divider formed by a resistor 182 and 184 extends between the conductors 152 and 154 so that the voltage between the conductors is measured. Typically, resistor 184 is much smaller than resistor 182. The voltage across resistor 184 is digitized by an analog-to-digital converter 184. It is assumed that the transfer functions at the potential divider are known, but the teachings of WO2014 / 072733 can be used to determine the transfer function and are hereby incorporated by reference. Likewise, the transfer characteristic of the current transformer can be assumed to be known, but if it needs to be determined then the reader is referred to the teaching of WO2013 / 038176, the teachings of which are incorporated by reference.
The outputs of analog-to-digital converter 184 are a series of samples Vs. Assuming that the current samples and voltage samples relate to substantially the same point in time (ie, the time separation between them is zero or very small in comparison to the period of the mains waveform), then the power drawn by the load can be represented as follows:
A processor 190 receives the samples Is and Vs and can process them in order, among other things, to calculate the power consumed and to keep a sum of the energy consumed. The processor can also review the series of scans to provide other services, such as troubleshooting, excessive loads, evidence of the tampering, and so on, that might be of interest to a utility. The processor can output the result of its calculations by means of a user interface 192, for example in the form of a display and / or by means of wireless or wired data connections 194 and 196.
It can be seen by looking at a sinusoid that a phase measurement error corresponds to the displacement of the sinusoid in time. Thus, the sample Is in the digital domain for a pure sinusoid is a displaced version of what it should be, and if the phase measurement error is known, the sample can be shifted by an amount of time corresponding to the phase measurement error and then in the calculation the power set forth in Equation 5 can be used. Where the current signal is a superposition of sinusoids with different frequencies, the designer has either the choice of using just a single time shift to compensate for the most important component, or checking the phase error as a function of frequency, and then the individual one Extract contribution from one or more of the important frequency components, move them back in time to their correct positions, and then calculate the power consumption. If phase angle data is required, the phase angle can be determined by a phase detector circuit or using FFT or Geortzel algorithms. In fact, it can be seen from the reference to the generalized situation in FIG. 1 that both the current measurement signal and the voltage measurement signal can be subject to phase shifts. The teachings of the present disclosure can be used to apply correction for phase measurement errors and displacements to both the voltage and current measurements so that they are properly timed.
The phase error at a particular frequency can be checked by generating a measurement signal from the phase error measurement circuit 170 at that particular frequency in accordance with the teachings set forth above. FIG. 14 shows a flowchart for characterizing the phase error at multiple frequencies. The process starts at step 200. Control passes to step 210, where a counter / register is initialized to a value N in order to obtain the first frequency to be examined F ( N). From here control goes to step 220, in which the current modulation circuit is designed to deliver a modulated current at the frequency F (N), so that the result of such a modulation can be detected in the output sequence by the analog-to-digital converter 174 . Control transfers to step 230 where a test is made to see if the phase measurement error needs to be determined at other frequencies. If so, control passes to step 240 where the value N is changed to represent a different frequency and control returns to step 220 so that the phase measurement error is determined at a different frequency. If step 230 determines that no other phase error measurement is needed, control passes to step 250 which waits until another phase measurement error update is scheduled before control passes back to step 210.
The phase measurement error estimates can be used to correct phase measurements immediately or stored for later use. Figure 15 is a flow diagram illustrating how the phase measurement errors can be used. Step 280 obtains a phase measurement error at step 280, for example from a value stored in memory as a result of executing the flowchart shown in FIG. 14, and at step 290 this is used to calculate a time correction necessary to shift the signal Is is used by S = TR, where T is the time from any system time where S = 0, and R is the sample rate to Is '= Is + (ϕR / (360F)), where Is' is a corrected sample number of the sample Is, ϕ is the phase measurement error in degrees, R is the sampling rate, and F is the frequency of the signal or signal component to which the correction is applied. This correction is applied at step 300. From step 300, the phase measurement error correction can optionally be carried out at other frequencies by setting a new frequency in step 310 and then returning the process sequence to step 280.
So far it has been assumed that the slew rate limited transitions are linear, but this need not be the case. Various distortions can affect the shape of an ideal square wave, an example of which is shown in Figure 16a.
A first form of distortion that has already been considered is slew rate limitation, in which the ideal immediate rising and falling transitions 320 and 322 of FIG. 16a are provided as slower transitions. Figure 16b illustrates exemplary shapes of rate-limited square waves in which the rising edge is represented by waveforms 330, 332, and 334 having fast, moderate, and slow slew rates, respectively. Likewise, the falling edge is represented by a relatively fast transition 340, an intermediate transition 342, and a relatively slow transition 344.
There is no reason to assume that the slew rates for the rising and falling transitions will be the same. Thus, the square wave-like input waveform can have an asymmetrical slew rate limited shape, as shown in Figure 16c.
Slew rate limiting is not the only form of distortion that could affect the input waveform. The on-resistance of transistors can be combined with parasitic capacitance to produce rising and falling edges that asymptote to their target value, as shown by junction 350 in Figure 16d, in the style of charging or discharging a capacitor through a resistor. Likewise, a parasitic inductance can interact with the parasitic capacitance, introducing overshoot, also shown in Figure 16d.
The presently disclosed embodiments of the invention can be used to estimate a corrected effective time of a rising edge and a falling edge for the square wave, with the timings being set at, for example, 50% of the voltage transition threshold. However, other values can also be selected.
As noted with reference to Figure 1, components such as the anti-aliasing filter, the ADC and the programmable gain amplifier can introduce a delay. This observation can be generalized further, as shown in FIG.
In FIG. 17, a transition with a rising edge is instructed by a digital instruction at time 400 and a transition with a falling edge is instructed at time 402. These instructions are provided to the signal generator 404. The signal generator can be a simple logic circuit, such as a D-type flip-flop, with the Q-bar output connected to its data input, or it could be more complex, such as a DAC. However, it can be assumed that the signal generator introduces a delay and has a limited transition speed, so the effective transitions should be placed at new times 410a and 412a. The output from the signal generator goes through a driver 420 which adds additional delay and / or slew rate and bandwidth limitation so the effective transitions should now be placed at times 410b and 412b. The current sensor 3 adds a further delay so that the effective transitions, as measured, are now at times 410c and 412c. By the time the signal has passed through filter 6, the effective transitions at times 410d and 412d have been shifted. By the time digitization has been completed at ADC 7, the effective transitions at times 410e and 412e have been delayed.
The relative magnitudes of any additional delay are purposely not drawn to scale. It should therefore simply be noted that in the current signal path and in the voltage signal path, each input signal that is used for measurement purposes can be subject to the sum of the delays, and that a correction may have to be applied to the voltage and current measurement channels.
As previously noted, the signal generator could be a DAC and therefore the input signal can be given any desired shape and since the shape of the input signal is known, the same shape can be searched for in the output of the ADC 7 to obtain an estimate of the To detect propagation delay. Thus, the DAC could be driven to generate classic waveforms such as square waves, incremental approximations of triangular waves, incremental approximations of sinusoids, and so on.
In an alternative approach, the DAC could also generate random or pseudo-random test sequences which would look like noise but which could still be recovered from the output of the ADC 7 to allow delay to be estimated. Autocorrelation techniques can be used to accomplish this because they are computationally robust and reasonably easy to perform. This would characterize the time delay through the system, which could then be converted to a phase delay for a given frequency.
The arrangement of FIG. 17, in which the signal generator 404 is a DAC, also opens up the possibility of using the DAC in order to generate a known approximation of a rise rate-limited square wave. However, now the transition rate of the rising and falling edges can be determined by the digital circuitry driving the DAC and these rates can be chosen to be comfortably within the bandwidth and slew rate capabilities of the buffer / driver 420. Now, the correction discussed hereinabove with reference to Figure 9 can be provided as a preset number based on the effective transition time, e.g. From T1 to T2 as defined by the control word sequence provided to the DAC. A similar deterministic, as opposed to a measurement-based, approach can be used for other signal profiles in which the characteristics of the signal output by the DAC with regard to the speed of voltage transitions are selected in such a way that they do not approach the accuracy limits of a downstream driver circuit that is required to introduce the input signal into the measurement circuit.
Such an observation can, however, be expanded further, as shown in FIG.
Here a signal can be generated from a signal source that we do not control in a deterministic way. The signal could come from a low quality oscillator and driver (including extremely low quality), a filtered noise source, or a random number generator driving a DAC. However, a copy of the reference / input stream is digitized by an analog-to-digital converter 450, which may be a separate device, or could be provided by the ADC 7, which operates in a time-division multiplexed manner, and the digitized input signal, that is used to characterize the CT response and the output from the CT can then be compared and cross-correlated to find the delay. Using the ADC 7 in a time division multiplexed manner to capture a copy of the input signal can be beneficial and the delay introduced by the filter 6 and the PGA / ADC 7 can be made common to both signal chains, effectively using the effects of these delays be weakened.
In the embodiments described with reference to FIGS. 7 and 8, a controller 60 supplied a timing signal to the power meter 50. The power meter 50 can, however, also be connected to an optional second current sink in order to provide a signal Iref, as shown in FIG , which accurately tracks when transistor 124 switches, thereby providing a direct measurement of the start of a transition.
The circuit can be used on a single phase, as shown in the figures, or on split-phase systems such as those in the United States or Japan, or on 3-phase systems, as commonly found in larger installations . In 3 phase systems, 3 current transformers would be used, one for each of the phases, and a neutral point may be connected to a return line to account for phase imbalance.
The circuit can be used in many applications where measurement of AC signals is desired and can be used in domestic, industrial, aeronautical, and medical fields (this is not a limiting listing). The apparatus and method described herein can be used "in situ," but they can also be used by component manufacturers and installers to perform testing and calibration during the manufacture and / or installation of transducers and measurement devices. The meter may have a communication capability (as is becoming common practice) to enable it to report power consumption. This ability can be effectively used to report the phase error, as well as for network monitoring purposes to identify uncompensated loads or to monitor the performance of the measuring devices, so that errors or a degradation in the performance can be identified and a correction and / or Compensation is planned or a correction is applied to a customer's invoice in order to avoid overcharging and thus intervention by a regulatory authority until repair or replacement of the measuring device and / or current transformer either alone or in combination.

权利要求:
Claims (30)
[1]
1. A method for estimating phase measurement errors in measurements of a measurand, the measurand being a current or a voltage, the method comprising:Providing an input signal with a fundamental frequency to an input of a measuring transducer (30; 160), the measuring transducer (30; 160) serving to measure the measured variable;Receiving an output signal from the transducer (30; 160);Analyzing the output signal to determine a phase difference compared to the input signal; andDetermining an estimated phase measurement error in the measurement of the measured variable by applying a first correction to the determined phase difference in order to compensate for a finite rate of change of the input signal.
[2]
2. The method of claim 1, wherein the input signal is a repetitive signal.
[3]
3. The method of claim 1, wherein the input signal is a step-wise approximation to a continuous signal.
[4]
4. The method of claim 3, wherein the continuous signal is either a sinusoid, a triangle wave, a square wave with smooth transitions, or a bandwidth limited noise source.
[5]
5. The method of claim 2, wherein the input signal is a square wave signal and the first correction comprises compensating for the finite rate of change of edges of the square wave signal as it transitions between high and low values.
[6]
6. The method according to any one of claims 2 or 5, wherein the repetitive input signal approximates a rise rate-limited square wave or a charge rate-limited square wave.
[7]
7. The method of claim 6, wherein the slew rate or charge rate limited square wave transitions between a first value (70) and a second value (72) andwherein applying the first correction comprises determining an estimate of a time to reach a midpoint between the first and second values (70, 72).
[8]
8. The method according to any one of claims 6 or 7, in which the time to reach the midpoint is formed by starting a counter or timer at the beginning of the transition and stopping it when the end point value is reached.
[9]
9. The method of claim 7, further comprising adding a second correction to account for non-linearity in the transition rate between the first value and the second value.
[10]
10. The method of claim 9, wherein the second correction is estimated or measured during manufacture of the transducer (30; 160) and is stored in a memory.
[11]
11. The method according to any one of the preceding claims, wherein the measured variable is a voltage, and wherein the measuring transducer (30; 160) comprises a voltage transducer.
[12]
12. The method of claim 11, wherein the voltage converter comprises a potential divider (182, 184).
[13]
13. The method according to any one of claims 1 to 10, wherein the measured variable is a current, and wherein the measuring transducer (30, 160) comprises a current transformer.
[14]
14. The method according to claim 13, wherein the measuring transducer (30, 160) is a current transformer and the phase difference after the application of the correction signal represents a phase shift that results from the current transformer (30; 160).
[15]
15. The method according to claim 6, wherein the increase rate or charge rate limited square wave has a substantially uniform mark-to-space ratio.
[16]
16. The method of claim 1, wherein the phase difference is estimated using an FFT or Goertzel algorithm or a phase detector circuit.
[17]
17. The method according to claim 1, wherein the input signal has a predetermined rise rate or transition time and is formed by a digital-to-analog converter (5; 174, 184, 450) and the first correction applied to compensate for the finite rate of change of the input signal is based on the predetermined slew rate or transition time of the input signal.
[18]
18. The method of claim 17, wherein the first correction is a preset number.
[19]
19. The method of claim 2, wherein the repetitive signal has nominally linear rising and falling edges, and whereina correction is applied to the estimated phase measurement error in order to compensate for a finite rate of change of the edges of the input signal,wherein the fundamental frequency of the input signal is adjustable so that the phase measurement error can be estimated at one or more frequencies.
[20]
20. A method for estimating a power consumption which comprises measuring the potential on a first conductor (20), measuring the current flowing in the first conductor (20), applying a phase correction based on a phase correction to the measurement of the current or the potential an estimated phase measurement error determined using the method of claim 1 and multiplying the corrected potential and current measurements to estimate the performance.
[21]
21. The method according to claim 20, wherein the correction to the measured current or to the measured voltage:comprises estimating and applying a time shift to be applied to sampled values of the voltage compared to sampled values of the current; orcomprises estimating and applying a time shift to be applied to sampled values of the current compared to sampled values of the voltage.
[22]
22. Apparatus for estimating phase measurement errors in measurements of measured quantities comprising a current or a voltage, the apparatus comprising:a signal generator (404) for providing an input signal to an input of a measuring transducer (30; 160), the measuring transducer (30; 160) serving to measure the measured variable; anda phase or time shift comparator for receiving an output signal from the transducer (30; 160) and analyzing the output signal to determine a phase or time difference compared to the input signal;wherein the phase or time shift comparator is designed to determine an estimated phase shift by applying a phase correction to the determined phase or time difference in order to compensate for imperfection artifacts in the input signal due to a finite rate of change of the input signal.
[23]
23. The apparatus of claim 22, further comprising circuitry (50) for measuring the transition time in the input signal due to a finite rate of change of the input signal.
[24]
24. Device according to one of claims 22 or 23, in which the signal generator (404) is a square wave generator.
[25]
25. Device according to one of claims 22 to 24, in which the measuring circuit (50) estimates the midpoint of the square wave transition and provides a timing signal to the phase corrector.
[26]
26. The device according to claim 22, wherein the input signal is an approximation of a square wave, but has ramp-like transitions, the device further comprising a digital-to-analog converter (5; 174, 184, 450) which is adapted to the To form input signal, so that the duration of the ramp-like transitions are known or predetermined and a known or predetermined correction value takes into account the duration of the ramp-like transition.
[27]
27. A power meter (50; 150) adapted to estimate the amount of power that is consumed by or transferred to an electricity distribution system, the power meter comprising a measuring circuit for measuring the power and an apparatus according to any one of claims 21 to 25 having.
[28]
28. The power meter (50; 150) of claim 27, further comprising an interface for sending data back to a network operator, the data being estimates of the performance of the power meter (50; 150) and / or information about load and voltage conditions at the power meter ( 50; 150).
[29]
29. Power meter (50; 150) according to claim 27, which is further designed to take harmonic signals into account in the measured variable when the drawn power is calculated.
[30]
30. Power meter (50; 150) according to claim 27, wherein the signal generator (404) generates a repetitive signal with a fundamental frequency and the frequency of the signal generator (404) is adjustable.

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同族专利:

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US10132846B2|2018-11-20|

JP2017223642A|2017-12-21|

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JP2021121802A|2021-08-26|

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JP6882542B2|2021-06-02|

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WO2017216598A1|2017-12-21|

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法律状态:
2020-11-13| AZW| Rejection (application)|

2021-01-15| AEN| Modification of the scope of the patent|Free format text: :DIE PATENTANMELDUNG WURDE AUFGRUND DES WEITERBEHANDLUNGSANTRAGS VOM 13.01.2021 REAKTIVIERT. |

优先权:

申请号 | 申请日 | 专利标题

US15/182,094|US10132846B2|2016-06-14|2016-06-14|Method of and apparatus for learning the phase error or timing delays within a current transducer and power measurement apparatus including current transducer error correction|

PCT/IB2016/001992|WO2017216598A1|2016-06-14|2016-12-21|Method of and apparatus for learning the phase error or timing delays within a current transducer and power measurement apparatus including current transducer error correction|

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