巴西专利BR102014001743B1 energy conversion system and method for controlling a multiphase energy conversion system

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专利摘要:
APPARATUS AND METHOD FOR CONTROLLING THE CURRENT CURRENT IN AN INVERTER SYSTEM. The present invention relates to an energy conversion system that provides multiphase energy, including phase voltages for each phase of multiphase energy. The system comprises a plurality of inverters that generate PWM output voltages based on PWM control signals. A plurality of inductive components are configured to receive the PWM output voltages to generate the phase voltages. PWM output voltages cause streams of circulating current through the inductive components. A voltage controller is employed, which is responsive to the phase voltages to generate voltage modulation signals that correspond to the phase voltages. A plurality of current sharing channels are respectively associated with each component of the plurality of inductive components and they are configured to generate current sharing modulation signals in response to circulating current flows. The PWM control signals are generated based on the modulation signals obtained by combining the current sharing modulation signals and the voltage modulation signals.
公开号:BR102014001743B1
申请号:R102014001743-7
申请日:2014-01-24
公开日:2021-02-09
发明作者:Qingquan Tang；Dazhong Gu；Dariusz Czarkowski；Francisco Leon；Kamiar J. Karimi；Shengyi Liu
申请人:The Boeing Company；
IPC主号:

专利说明:

BACKGROUND
[001] Energy converters are used in electrical power systems for aircraft, as well as power systems for other devices. The electrical power systems in today's commercial airplanes are mainly provided by 400 Hz, 115 V or 230 V three-phase AC power supplies. The power system can include one or more alternative low-voltage DC power supplies, such as a battery of fuel cells or a battery, which supplies the input power to a pulse width modulated (PWM) energy conversion system. The outputs of multiphase voltages, such as three-phase voltages, can be provided to an aircraft electrical power distribution system, which supplies electrical power to a downstream distribution system. The downstream distribution system can have loads of various types, including, but not limited to, a three-phase, single-phase conversion system or another with DC loads, etc.
[002] Many power converters, however, are not fully optimized for aircraft applications. Such energy converters can be large and heavy, increasing the weight of the plane and limiting the volume available to the other components of the plane. To resolve this issue, power converters can include parallel or interleaved inverters. With the use of inverters in parallel or interleaved, the conversion systems can achieve a higher energy while simultaneously using devices of lower nominal regime, thus also providing a higher efficiency, a higher energy density (measured in kW / kg), and savings in weight and volume. In addition, interleaved converters improve harmonic reduction compared to non-interleaved converters. However, such converter systems can generate circulating current, which can degrade performance or cause malfunctions, even to the point of damaging the user's equipment connected to the power bus.
[003] In power converters that use inverters in parallel or interleaved, the inverter outputs can be connected to inductive components to limit the circulating current. However, inductive components often do not work well in the low-frequency circulating current. Low-frequency circulating currents can cause saturation of the cores of inductive components. The saturation of the cores can reduce the performance of the energy converter, as well as disable the conversion system.
[004] Also in the design of the energy converter, large magnetization inductances may be desired to reduce core loss and better limit high-frequency circulating currents. However, this may require advanced and accurate knowledge of system parameters, which makes the design process complicated and time-consuming. For example, the complexity of the design of the control system can be caused by a reduced margin in the flow of a given magnetic core when a large magnetization inductance is desired.
[005] Therefore, there are at least two problems associated with energy conversion systems. They may experience reduced performance when used with high transient loads. In addition, the design process can be complicated and time-consuming. SUMMARY
[006] An energy conversion system is provided that provides multiphase energy, including phase voltages for each phase of multiphase energy. The system comprises a plurality of inverters that generate PWM output voltages based on PWM control signals. A plurality of inductive components are configured to receive the PWM output voltages to generate the phase voltages. PWM output voltages cause streams of circulating current through the inductive components. A voltage controller is employed, which is responsive to the phase voltages to generate the voltage modulation signals that correspond to the phase voltages. A plurality of current sharing channels are associated with each component of the plurality of inductive components respectively and they are configured to generate current sharing modulation signals in response to circulating current flows. The PWM control signals are generated based on the modulation signals obtained by combining the current sharing modulation signals and the voltage modulation signals.
[007] In addition, the invention comprises modalities according to the following clauses:
[008] Clause 1. An energy conversion system configured to supply multiphase energy that has a phase voltage associated with each phase of the multiphase energy, respectively, in which the energy conversion system comprises:
[009] a plurality of inverters configured to generate PWM output voltages for each phase voltage in response to PWM control signals;
[0010] a plurality of inductive components configured to receive the PWM output voltages to generate the phase voltages, wherein the PWM output voltages result in a plurality of circulating current flows in the plurality of inductive components;
[0011] a responsive voltage controller to generate voltage modulation signals for each phase voltage;
[0012] a plurality of current sharing channels associated with each component of the plurality of components respectively, wherein the plurality of current sharing channels is configured to generate current sharing modulation signals in response to the plurality of current flows current; and
[0013] a carrier reference circuit configured to generate the PWM control signals in response to the modulation signals, in which the modulation signals are obtained by combining the current sharing modulation signals and the voltage modulation signals.
[0014] Clause 2. The energy conversion system according to clause 1, in which the inductive components consist of a plurality of transformers between cells.
[0015] Clause 3. The energy conversion system according to clause 1 or 2, in which the inductive components consist of a plurality of inductors.
[0016] Clause 4. The energy conversion system according to clause 2, in which the plurality of inverters comprise first and second inverters, and in which the first and second inverters generate three-phase PWM output voltages interspersed.
[0017] Clause 5. The energy conversion system according to clause 4, in which the plurality of transformers between cells is configured to supply three-phase supply voltages when using interleaved three-phase PWM output voltages of the first and second inverters.
[0018] Clause 6. The energy conversion system according to any one of clauses 1 to 5, in which each of the first and the second inverters also provides a neutral phase voltage.
[0019] Clause 7. The energy conversion system according to any of the preceding clauses, which also comprises a current detection circuit that has a plurality of current sensors configured to supply signals from the circulating current to the sharing channels. current for each component of the plurality of inductive components.
[0020] Clause 8. The energy conversion system according to any of the preceding clauses, in which the current sharing channels comprise a resonant fundamental frequency controller that has a central frequency close to the fundamental frequency of the phase voltages .
[0021] Clause 9. The energy conversion system in accordance with any of the preceding clauses, in which the current sharing channels have a frequency transformation response that generally corresponds to:
DC Low frequency Fundamental frequency where

[0022] where wi defines a center frequency for a low frequency filter, Wf defines a center frequency for a fundamental frequency filter, Δwl defines a bandwidth for the low frequency filter, Δwf defines a bandwidth for the low frequency filter fundamental frequency, Kl0 and Kf0 define the pass band magnitudes of the low frequency filter and the fundamental frequency filter, respectively, Kl and Kf define peak gains of the low frequency filter and the fundamental frequency filter, respectively, and Cph ( s) is a phase delay compensator that provides phase compensation around the fundamental frequency.
[0023] Clause 10. The energy conversion system according to any of the preceding clauses, which also comprises a DC power source configured to supply input power to the plurality of inverters, in which the plurality of inverters share a bus of common CC.
[0024] Clause 11. A control system for an energy conversion system, comprising:
[0025] a voltage controller responsive to three-phase voltage signals to generate the corresponding voltage modulation signals;
[0026] current sharing channels responsive to a plurality of circulating currents of a plurality of inductive components, in which the inductive components are associated with the respective phases of the three-phase voltage signals, in which the current-sharing channels generate the signals of current sharing modulation associated with each phase of the three-phase voltage signals respectively when using the plurality of circulating currents of the plurality of inductive components; and
[0027] a plurality of combining circuits, in which each combining circuit is associated with a phase of the three-phase voltage signals, and in which the combining circuits combine the voltage modulation signals and the current sharing modulation signals associated respectively with each phase of the three-phase voltage signals to generate PWM control signals.
[0028] Clause 12. The control system according to clause 11, in which the current sharing channels are configured to receive a plurality of circulating currents from transformers between cells.
[0029] Clause 13. The control system according to clause 11 or 12, in which the current sharing channels are configured to receive a plurality of inductor circulating currents.
[0030] Clause 14. The control system according to clause 12, in which the current sharing channels comprise a resonant frequency controller that has a central frequency close to the fundamental frequency of each phase of the three-phase voltage signals.
[0031] Clause 15. The control system according to clause 13, in which each current sharing channel has a frequency transformation response that generally corresponds to:
DC Low frequency Fundamental frequency where

[0032] where wi defines a center frequency for a low frequency filter, wf defines a center frequency for a fundamental frequency filter, Δwl defines a bandwidth for the low frequency filter, Δwf defines a bandwidth for the low frequency filter fundamental frequency, Kl0 and Kf0 define the pass band magnitudes of the low frequency filter and the fundamental frequency filter, respectively, Kl and Kf define peak gains of the low frequency filter and the fundamental frequency filter, respectively, and Cph ( s) is a phase delay compensator that provides phase compensation around the fundamental frequency.
[0033] Clause 16. An energy conversion system configured to supply multiphase energy, in which the multiphase energy has a phase voltage associated with each phase of multiphase energy, respectively, in which the energy conversion system comprises:
[0034] a PWM activation circuit;
[0035] a plurality of inverters that have inputs coupled to the PWM activation circuit;
[0036] a plurality of inductive components coupled to the PWM voltage outputs of the plurality of inverters;
[0037] a plurality of power supply output terminals coupled to the outputs of the plurality of inductive components;
[0038] a plurality of current sensors coupled to the plurality of inductive components;
[0039] a voltage controller coupled to the plurality of inductive components;
[0040] a current sharing system coupled to the plurality of current sensors;
[0041] a combining circuit coupled to the outputs of the voltage controller and to the outputs of the current sharing system; and
[0042] an activation circuit coupled to the combiner circuit.
[0043] Clause 17. The energy conversion system according to clause 16, in which the plurality of inductive components includes transformers between cells.
[0044] Clause 18. The energy conversion system according to clause 16, in which the plurality of inductive components includes inductors.
[0045] Clause 19. The energy conversion system according to clause 17, in which the plurality of current sensors is coupled to transformers between cells to provide signals that correspond to twice the circulating current that flows through each one of the transformers between cells.
[0046] Clause 20. The energy conversion system according to any of clauses 16 to 19, wherein the current sharing system comprises a plurality of current sharing controllers arranged in pairs, with each pair of the plurality current-sharing controllers is coupled to a respective current sensor.
[0047] Clause 21. The energy conversion system according to clause 20, in which each current sharing controller of each pair of current sharing controllers comprises a resonant controller that has a central frequency close to a fundamental frequency voltage in the plurality of power supply terminals.
[0048] Clause 22. The energy conversion system according to any one of clauses 16 to 21, which also comprises a DC power supply coupled to the plurality of inverters, in which the plurality of inverters is coupled to a busbar. Common CC.
[0049] Clause 23. A method for controlling a multiphase energy conversion system, comprising:
[0050] the generation of voltage modulation signals in response to each phase voltage of the multiphase energy;
[0051] the detection of circulating currents that flow through a plurality of transformers between cells, each of which is associated with a respective phase voltage;
[0052] the application of a controller to the circulating currents detected in accordance with the generation of current sharing modulation signals that correspond to the circulating currents; and
[0053] the combination of voltage modulation signals and current sharing modulation signals to generate PWM control signals.
[0054] Clause 24. The method according to clause 23, in which the application of the controller comprises the application of a resonant controller at a central frequency close to a fundamental frequency of each phase of the multiphase energy.
[0055] Clause 25. The method according to clause 23, in which the application of the controller comprises the application of a controller that has a transformation frequency response that in general corresponds to:
DC Low frequency Fundamental frequency where

[0056] where wi defines a center frequency for a low frequency filter, wf defines a center frequency for a fundamental frequency filter, Δwl defines a bandwidth for the low frequency filter, Δwf defines a bandwidth for the low frequency filter fundamental frequency, Kl0 and Kf0 define the pass band magnitudes of the low frequency filter and the fundamental frequency filter, respectively, Kl and Kf define peak gains of the low frequency filter and the fundamental frequency filter, respectively, and Cph ( s) is a phase delay compensator that provides phase compensation around the fundamental frequency.
[0057] The characteristics, functions and advantages that have been discussed can be obtained independently in several modalities, or can be combined in other modalities whose additional details can be seen by reference to the description below and the drawings. BRIEF DESCRIPTION OF THE DRAWINGS
[0058] Figure 1 is a block diagram of an energy conversion system that uses transformers between cells as inductive components.
[0059] Figure 2 illustrates a way in which the transformers between cells in Figure 1 can be coupled within the energy conversion system.
[0060] Figure 3 illustrates a voltage control system that can be used within the energy conversion system of Figure 1.
[0061] Figure 4 illustrates a way to perform sequence decomposition.
[0062] Figure 5 illustrates an algorithm configured to perform the sequence decomposition shown in Figure 4.
[0063] Figure 6 illustrates the current sharing channels that can be used in the energy conversion system of Figure 1.
[0064] Figures 7a - 7c are Bode diagrams for an example of a current sharing controller.
[0065] Figure 8 illustrates an energy conversion system in which the various signal processing operations take place in a digital signal processor (DSP).
[0066] Figure 9 shows a method for controlling an energy conversion system.
[0067] Figures 10a - 10b are graphs of exemplary signals associated with the voltages of an energy conversion system that does not implement a control system of the type shown and described with respect to one or more Figures 1 to 9.
[0068] Figures 11a - 11b are graphs of exemplary signals associated with the voltages of an energy conversion system that has a control system of the type shown and described with respect to one or more Figures 1 to 9.
[0069] Figure 12 is a flow chart that illustrates how the energy conversion system can be incorporated in the context of the design and operation of an airplane.
[0070] Figure 13 is a block diagram that illustrates an airplane that incorporates the energy conversion system. DESCRIPTION
[0071] Figure 1 is a block diagram of an energy conversion system 100. The energy conversion system 100 includes an energy activation section 105 and a control system 110. The energy conversion system 100 provides multiphase energy at load 115. Although the following modalities are described in the context of a three-phase inverter system that provides three voltages in phase at about 120 ° from each other, the modalities can be extended to inverter systems that have more or less than three phases. EXAMPLE ENERGY ACTIVATION SECTION
[0072] The power activation section 105 includes a plurality of inverters, each of which has a plurality of PWM output voltages. The number of PWM output voltages provided by each drive is at least as large as the number of phases used to activate load 115.
[0073] In the exemplary energy conversion system 1000 of Figure 1, the energy activation section 105 includes a first inverter 120 and a second inverter 125, which receive DC power from a DC source 127. The first inverter 120 and the second inverter 125 shares a common DC bus from the DC source 127.
[0074] The first inverter 120 provides a first output voltage of PWM Vinva1, a second output voltage of PWM Vinvb1 and a third output voltage of PWM Vinvc1. Similarly, the second inverter 125 provides a first PWM Vinva2 output, a second PWM Vinvb2 output voltage and a third PWM Vinvc2 output voltage. The output voltages of PWM are interleaved.
[0075] The energy conversion system 100 in the example is configured as a four leg system. In this way, each of the first inverter 120 and the second inverter 125 includes neutral PWM output voltages. More particularly, the first inverter 120 provides a first neutral PWM output voltage Vinvn1, and the second inverter 125 provides a second neutral PWM output voltage Vinvn2. Such three-legged four-leg inverters can be used to maintain a desired sine waveform of the output voltage at each phase output over a desired range of load and transient conditions. The energy conversion system 100 does not need to be configured as such a four-legged system, but will be discussed in the context of such an architecture.
[0076] The energy activation section 105 also includes a plurality of inductive components. The inductive components can be in the form of inductors or transformers between cells. For purposes of describing the exemplary energy conversion system 100, transformers between cells are used as inductive components. However, the transformers between cells in Figure 1 can be replaced by inductors depending on the parameters of the system design.
[0077] In Figure 1, a first transformer between cells 175 is coupled to receive the first PWM output voltage Vinva1 from the first inverter 120 and the first PWM output voltage Vinva2 from the second inverter 125. A second transformer between cells 180 is coupled to receive the second PWM output voltage Vinvb1 and the second PWM output voltage Vinvb2. A third transformer between cells 185 is coupled to receive the third PWM Vinvc2 output voltage and the third PWM Vinvc2 output voltage. A fourth transformer between cells 190 is coupled to receive the neutral PWM output voltage Vinvn1 and the neutral PWM output voltage Vinvn2.
[0078] Figure 2 shows a way in which transformers between cells can be coupled to a power activation section 105. As shown, the output terminals of the first inverter 120 and the second inverter 125 are connected to the respective terminals of the first transformer between cells 175, second transformer between cells 180, third transformer between cells 185, and fourth transformer between cells 190. The transformer points between cells show the coupling configuration of the transformer windings. The cell-to-cell transformers in this example are configured as differential mode inductors. Although the magnetization inductance is used to limit the circulating current, the leakage inductance is used as an inductance for an output LC filter associated with each voltage phase respectively. When transformers between cells use a high permeability core, a high magnetization inductance is obtained. In this way, a small circulating current at high efficiency can be obtained.
[0079] Returning to Figure 1, the parallel operation of inverters 120 and 125 results in circulating currents in each transformer between cells. In the example, the current flowing through the first transformer between cells 175 is the difference in currents (Ia1 - Ia2) between the terminal containing the first PWM output voltage Vinva1 and the terminal containing the first PWM output voltage Vinva2. The current flowing through the second transformer between cells 180 is the difference in currents (Ib1 - Ib2) between the terminal containing the second PWM output voltage Vinvb1 and the terminal containing the second PWM output voltage Vinvb2. The current flowing through the third transformer between cells 185 is the difference in currents (Ic1 - Ic2) between the terminal containing the third PWM output voltage Vinvc1 and the terminal containing the third PWM output voltage Vinvc2. The current flowing through the fourth transformer between cells 190 is the difference in currents (In1 - In2) between the terminal containing the neutral PWM output voltage Vinvn1 and the terminal containing the neutral PWM output voltage Vinvn2.
[0080] The outputs of transformers between cells are provided with a current detection circuit 195. Although current detection circuit 195 is shown on the outputs of transformers between cells, it can alternatively be placed to monitor the current at the inputs of transformers between cells.
The current detection circuit 195 may include a plurality of current sensors, each of which is respectively associated with a voltage phase. Here, each transformer between cells includes two output terminals. The two output terminals of each transformer between cells are coupled to a respective Hall effect current sensor before fusing through the Hall effect current sensor in reverse directions at the coupled nodes to supply multiphase energy to the load. In this way, the difference in currents, or the circulating current, between the two output currents of each transformer between cells is acquired.
[0082] In Figure 1 and Figure 2, a first current sensor 200 is coupled to the output terminals of the first transformer between cells 175, where a first VA phase supply voltage is provided to load 115 at node 205. A second current sensor 210 is coupled to the output terminals of the second transformer between cells 180, where a second phase voltage supply VB is provided to load 115 at node 215. A third current sensor 220 is coupled to the output terminals of the third transformer between cells 185, where a third phase voltage supply VC is provided to the load 115 at node 225. A fourth current sensor 230 is coupled to the output terminal of the fourth transformer between cells 190, where a neutral phase voltage Vn is provided to load 115 at node 235. In this way, three-phase supply voltages (VA, VB, VC) are provided at load 115.
[0083] Transformers between cells suppress the high frequency circulating current. The low frequency circulating current passes through each transformer between cells and is detected by the current current detection circuit 195 for the control of the low frequency circulating current.
[0084] A capacitor is coupled to each node that contains a voltage for load 115. The respective capacitor for each voltage supply phase and the corresponding transformer inductance between cells can be used as a filter for the voltage supply phase. . In Figure 2, a first capacitor 240 is coupled to node 205 and forms a filter with the leakage inductance of the first transformer between cells 175 and the fourth transformer between cells 190 to filter the VA output phase voltage. A second capacitor 245 is coupled to node 215 and forms a filter with the leakage inductance of the second transformer between cells 180 and the fourth transformer between cells 190 to filter the VB output phase voltage. A third capacitor 250 is coupled to node 225 and forms a filter with the leakage inductance of the third transformer between cells 185 and the fourth transformer between cells 190 to filter the VC output phase voltage.
[0085] The power activation section 105 can also include a voltage detection circuit 263. As shown in Figure 2, the voltage detection circuit 263 includes a plurality of voltage dividers placed through capacitors 240, 245 and 250 to monitor the magnitudes of the VA, VB and VC supply voltages with respect to Vn. The divided voltage signals are provided to the power control system 110. In this example, the voltage divider includes the resistors connected through each capacitor.
[0086] The energy activation section 105 may include the low-pass filters associated with the current detection circuit 195 and the voltage detection circuit 263 respectively. In the example in Figure 1, the low-pass filters 300 are coupled to receive signals from the current detection circuit 195 along the current detection bus 305. The low-pass filters 310 receive voltages from the voltage detection circuit 263 and provide the output voltages Van, Vbn and Vcn along the current bus. voltage detection 315. CONTROL SYSTEM
[0087] With reference again to Figure 1, the control system 110 is coupled to receive the voltage signals Van, Vbn, Vcn on the voltage detection bus 315 for the provision of a voltage control system 325. The system control unit 110 is also coupled to receive the signals from the current detection circuit 195 to provide a plurality of current sharing channels 330. The outputs of the current sharing channels 330 and the outputs of the voltage control system 325 are provided to a plurality of combiners 340. The outputs of the combiners 340 are modulation signals that are obtained by combining the modulation current sharing signals generated by the current sharing channel 330 with the voltage modulation signals generated by the control system. voltage control 325. These modulation signals are provided for comparison with reference carrier signals in carrier reference 345 and carrier reference 350. Bearer reference 345 generates PWM control signals 355 for gate activators 360, which provide gate enable signals 365 to first drive 120. Similarly, bearer reference 350 generates PWM control 370 for gate activators 360, which provide gate 375 activation signals to the second inverter 125. EXEMPLIFIER VOLTAGE CONTROL SYSTEM
[0088] An exemplary voltage control system 325 is shown in Figure 3. In this example, the voltage signals Van, Vbn, Vcn are provided to the analog to digital converter 398, which converts the received voltage signals into the digital signals va , vb and vc on the digital signal bus 327. The digital signals on the digital signal bus 327 are provided to a sequence decomposer 400. The signals generated by the sequence decompositor 400 are provided at the input of an abc transformer on dq 405. The abc transformer on dq 405 transforms digital signals at their inputs into digital signals that can be manipulated in a dq coordinate system. Such manipulations in this example are performed by a voltage controller 410, which receives the signals from the abc transformer at dq 405. The voltage controller 410 performs operations on the dq signals from the abc transformer at dq 405 to generate the corresponding dq output signals. for a dc transformer on abc 415. A voltage modulation signal determiner 420 operates on the abc signals of the dq transformer on abc 415 to generate the respective voltage modulation signals Vam, Vbm, Vcm, Vnm, on the signal bus digital 425. EXAMPLIFYING SEQUENCE DECOMPOSITION
[0089] As noted, the digital voltage signals on the digital signal bus 327 are broken down into positive, negative, and zero sequences by the sequence decomposer 400. If load 115 is unbalanced, the three-phase voltage and current may fluctuate in the system of coordinates dq. Therefore, it may be desirable to decompose the unbalanced voltage and / or current in three symmetrical three-phase systems. A general example of how this can be done in any generic three-phase system is illustrated by the following equations:

[0090] where (XA> P, xBp, xCp) is the vector of the positive sequence for the three-phase voltage and / or the current output, (XA „, XB„, xCpt) is the vector of the negative sequence, and (xAh , xBh, XO) is the null sequence vector. The vector (, XB, XC) corresponds to the vector of the three-phase voltage and / or current.
[0091] The positive, negative and null strings can be obtained by using the following equations:

[0092] where a = eJ2π / 3.
[0093] Supposing
then
. This sequence decomposition is illustrated graphically in Figure 4 and is applicable to the 100 energy conversion system.
[0094] To obtain the shape of the voltage and / or current vector, the imaginary part of the vector is obtained by executing a quarter of a fundamental cycle delay on the time domain signals of the three-phase voltage and / or current. A block diagram showing an implementation of such a sequence decomposition algorithm configured to perform the mathematical operations above is illustrated in Figure 5. EXAMPLE OF ABC / DQ TRANSFORMATIONS
[0095] A direct quadrature - zero (dq) transformation is a mathematical transformation used to simplify the analysis of three-phase circuits. With balanced three-phase circuits, the application of dq transformation reduces the three quantities of AC to two quantities of DC. Simplified calculations can then be performed on these imaginary quantities of DC before performing the reverse transformation to recover the modified three-phase AC results. In this way, the dq transformation operations can simplify the calculations performed by the 325 voltage control system.
[0096] An example of a transformation from dq to a three-phase voltage is shown here in the form of a matrix:

[0097] This transformation is performed by the abc transformer at dq 405 at the received voltages. An inverse of this transformation is performed by the dq transformer in abc 415. The inverse transformation is:
EXEMPLIFIER VOLTAGE CONTROLLER
[0098] The voltage controller 410 can perform proportional - integral (PI) operations on the dq signals received from the abc transformer at 405 dq. For this purpose, the voltage controller 410 may include a PI controller that has the following frequency response:

[0099] In certain applications, the PI controller can be modified to meet the stability and dynamic response requirements of the system. In this way, a "two pole controller" that has two poles can be used. More particularly, the two-pole controller can have the following frequency response:

[00100] Such a two-pole controller can provide a greater bandwidth and a greater magnitude margin for the voltage controller 410 than the PI controller in the first example.
[00101] In this two-pole controller, Mv2 is selected below the overflow frequency of the Bode diagram between the system voltage and the control magnitude, to provide a high damping, thus ensuring a high magnitude margin for the system. The value for Mv1 is selected to obtain the desired phase margin of the voltage and control system (60 degrees in the three-phase system described here), and Kv is selected as an option between the robustness of the system and the bandwidth (response speed) ). The values for Kp and Ki determined the gain and zero of the transfer function. The gain is selected as an option between the robustness of the system and the bandwidth (response speed). Zero is selected to obtain the desired phase margin. EXAMPLE OF CHAIN SHARING CHANNEL
[00102] An example of a current sharing channel 330 is illustrated in Figure 6. As shown, four differential digital circulating current signals are received on separate lines from the 413 bus. Each differential circulating current is associated with each transformer respectively. cells and provided with a respective current sharing channel 330. For the sake of simplicity, only the current sharing channel 330 associated with the circulating current (Ia1 - Ia2) of the first transformer 175 between cells is described. The remaining current sharing channels 330 associated with the second transformer between cells 180, the third transformer between cells 185 and the fourth transformer between cells 190 have the same structure. Two or more current sharing channels 330 for different voltage phases can operate in parallel in a generally concurrent manner.
[00103] The digital signals on bus 413 that correspond to the signals of circulating currents (Ia1 - Ia2) are provided at the input of a first amplifier 430 and at the input of a second amplifier 435. The first amplifier 430 multiplies the signals of circulating currents by a factor of -0.5, while the second amplifier 435 multiplies the signals of circulating currents by a factor of +0.05. The output of the first amplifier 430 is provided to the input of a first current sharing controller 440, and the output of the second amplifier 435 is provided to an input of a second current sharing controller 445. Output 450 of the first sharing controller current 440 is provided to an input of a first combiner 455, and output 460 of the second current sharing controller 445 is provided to an input of a second combiner 465. The signal at output 450 corresponds to a current modulation signal as generated by the current sharing controller 440. The signal at output 460 corresponds to a current modulation signal as generated by the second current sharing controller 445. Amplifiers 430 and 435 place the current modulation signals one off with others.
[00104] In addition to the current modulation signals, each current sharing channel 330 receives a respective voltage modulation signal for a given three-phase voltage phase of the 425 bus. Regarding the circulating current (Ia1 - Ia2) of the first transformer between cells 175, the corresponding Vam voltage modulation signal is provided to an input of the first combiner 455 and to an input of the second combiner 465. The first combiner 455 provides a first modulation signal at output 470 which corresponds to a sum the current modulation signal generated by the current sharing controller 440 and the Vam voltage modulation signal generated by the voltage control system 325. The second combiner 465 generates a second modulation signal at output 475 which corresponds to a sum of the current modulation signal generated by the second current sharing controller 445 and the Vam voltage modulation signal generated by the control system voltage control 325. The modulation signal at output 470 can be provided to carrier reference circuit 345 for comparison with a corresponding carrier signal to generate the PWM 355 control signals used to control the first PWM output voltage Vinva1 of the first inverter 120. The modulation signal at output 475 can be provided to carrier reference circuit 350 for comparison with a corresponding carrier signal to generate the PWM control signals 370 used to control the first output voltage of Vinva2 PWM of the second inverter 125. In each example, the PWM control signals are provided to the door activators 360 for the respective inverters. EXAMPLE OF CURRENT SHARING CONTROLLER
[00105] An example of a structure for a 440 current sharing controller (C (s)) is exemplified in the following equation:
DC Low frequency Fundamental frequency where

[00106] Here, wi defines a central frequency of a low frequency resonant filter, wf defines a central frequency of a resonant fundamental frequency filter, Δwl defines a bandwidth of the low resonant filter, Δwf defines a bandwidth of the resonant fundamental frequency filter, Kl0 and Kf0 define the pass band magnitudes of the resonant low frequency filter and the fundamental frequency resonant filter, respectively. Kl and Kf define peak gains of the low resonant frequency filter and the fundamental resonant frequency filter, respectively, and Cph (s) is a phase delay compensator that provides phase compensation around the fundamental frequency. The resonant fundamental frequency controller has a central frequency close to a fundamental frequency of the supply voltage for each phase of the multiphase (three-phase) voltage. It can also be seen that each current sharing channel 330 has the same frequency response vis-à-vis the respective current sharing controllers.
[00107] The values for kp and ki determine the gain and zero of the transfer function. Values are selected based on the desired robustness of the system. The values of kp and ki are selected to ensure a low cutoff frequency of the "CC" part of C (s) to obtain the desired robustness of the system.
[00108] In a three-phase power system that operates at 400 Hz, the value of 400 Hz is assigned as the value of Wf, which corresponds to the fundamental frequency. The value for w is selected so that it is at a low frequency, such as within a range of about 1 to 20 Hz. The value for ΔWl must be a relatively large number compared to ΔWf, which must be a small number . The values for Kl0 and Kf0 are selected to obtain a unit gain in the non-pass frequency range for the "low frequency" and "fundamental frequency" parts of C (s). The values for Kl and Kf are selected to obtain high peak values at the central frequency of the low frequency and fundamental frequency resonant controller, whereas the effect of ΔWl and ΔWf, Kl0 and Kf0 in these values can also be considered. For this purpose, increased Kl and Kf will have an effect similar to the increase in ΔWl and ΔWf (increasing the passband width of the low frequency and fundamental frequency resonant controllers), or increased Kl0 and Kf0 (increasing the gain in the range that does not pass through the low frequency and fundamental frequency resonant controllers).
[00109] The parameters of the phase delay compensator Cph (s) are selected based on the phase delay caused by the current detection circuit. For example, a time delay of 10-100 microseconds can be caused by the current detection circuit, which is equal to 1.44 ° - 14.4 ° at a fundamental frequency of 400 Hz. The Cph phase delay compensator (s) thereby compensates for a phase delay of 20-30 degrees at the fundamental frequency Wf assists to ensure system stability.
[00110] The low cutoff frequency of the "CC" part of C (s) helps in providing the stability of the system. The Wl that defines the center frequency of the low frequency resonant filter of C (s) can be selected so that it is in a range between the cutoff frequency of the "DC" part and the 400 Hz value of Wf. For example, Wl may be in the range of 1 to 20 Hz, with the passage bandwidth in the range of about 10 Hz to 30 Hz. The center frequency of the "fundamental frequency" part of C (s), as noted above, it is at 400 Hz, and may have a very small passage bandwidth. The phase angle of the phase delay compensator Cph (s) at 400 Hz should be selected to compensate for the time delay caused by the current detection circuit, and the magnitude before the cutoff frequency should be as close to the unit as possible .
[00111] When using the guidelines above, the C (s) values for a modality of a three-phase system are:
Where

[00112] Figures 7a - 7c are Bode diagrams for the 440 current sharing controller (C (s)). Figure 7a shows the frequency and phase response associated with each current sharing controller, where upper diagram 485 is in the plane's coordinate system showing the magnitude frequency response of a current sharing controller, and the diagram 490 is the phase response of the current sharing controller 440. In this example, the frequency and phase response of the DC filter is shown in 495. The frequency and phase response of the low frequency resonant filter is shown in 500. The frequency and phase response of the resonant fundamental frequency filter is shown in 505.
[00113] Figure 7b shows the frequency and phase response associated with the phase delay compensator Cph. More particularly, upper diagram 510 shows the magnitude of the frequency response at 515, while diagram 520 shows the phase response at 525.
[00114] Figure 7c consists of diagrams showing the total composite frequency and phase responses of the current sharing controller, including that of the phase delay compensator. More particularly, the upper diagram 525 shows the composite magnitude frequency response, while the lower diagram 530 shows the composite phase response. As illustrated, there is a peak 535 in the response shown in diagram 525 to the fundamental frequency of the voltage signals used to activate the load. Here, current sharing controllers are designed for an airplane, so that the total peak response 535 occurs at a frequency of about 400 Hz. The composite phase also shows a peak phase shift 537 approaching the fundamental frequency. IMPLEMENTATION OF THE DIGITAL SIGNAL PROCESSOR (DSP)
[00115] Figure 8 illustrates a power conversion system 100 in which several signal processing operations take place on a DSP 600. In the power conversion system 100, the three-phase output voltage is supplied to the load in operation 605, and these output voltages are detected in voltage detection operation 610 and optional low-pass filtering can occur in filtering operation 615 before signals are provided to a DSP 600 analog to digital converter (not shown) for handling in the digital domain.
[00116] Once the detected voltages are converted into digital signals, they are subject to a 620 sequence decomposition operation. The 620 sequence decomposition operation includes dividing the digital signals into positive, negative and zero sequences. Such operations are described above with respect to the sequence decomposer 400 of Figure 3.
[00117] Each positive, negative and null sequence is subjected to transformations from abc to individual dq. In this example, positive sequences are subject to transformation operations performed on the abc transformer on dq 625. Negative sequences are subject to transformation operations performed on the abc transformer on dq 630. Null sequences are subject to transformation operations performed by the abc transformer on dq 635.
[00118] The d-axis and q-axis outputs of each abc transformer in dq 625, 630 and 635 are provided to two voltage controllers that operate in the dq domain. In the illustrated example, the dq signals are provided to the respective controllers with a plurality of two-pole controllers 640. The operations performed by the two-pole controllers 640 can be those described above in connection with the PI controllers used in the voltage controller 410 of Figure 3.
[00119] The outputs of the two-pole controllers 640 are subject to an operation of transforming dq into abc in 645. The resulting abc signals are used in connection with the generation of modulation signals for each voltage phase of the three voltage output power. The abc signals are provided directly to the supermodulation module 650 for the execution of a supermodulation technique. The supermodulation technique can be any one of several such techniques.
[00120] The abc transformation of the neutral leg voltage is provided to a neutral leg modulation signal generator 655 before being processed by the supermodulation module 650. The signals provided at the outputs of the supermodulation module 650 correspond to the modulation signals of the tension control system 325 described above in connection with Figures 1, 3 and 6.
[00121] A plurality of transformers between cells 660 is used to supply the three-phase output supply voltages to the load in response to the PWM energy signals received from the first inverter 120 and the second inverter 125. The signals corresponding to the circulating currents that flow through each transformer between cells from a plurality of transformers between cells 660 are on the current detection bus 305 for analog to digital conversion within the DSP 600. Circulating current detection can be performed in the manner shown in Figure 1 and Figure 2. The signals on the current detection bus 305 are optionally provided to the low-pass filters 300 before going through the analog to digital conversion within the DSP 600. Due to the fact that the circulating current contains twice the switching frequency signal and since the sampling frequency of digital controllers can be limited, control close to e sampling wind from the digital controller may be required. For example, sampling timing can be activated at the peaks of PWM carrier signals to avoid introducing a false fundamental frequency component into the sampled circulating current.
[00122] In Figure 8, only a single current sharing channel 330 is shown. However, the DSP 600 performs operations for a plurality of current sharing channels 330, each of which is respectively associated with at least one transformer between cells corresponding to the plurality of transformers between cells 660.
The current modulation signals are provided along a passage 670 to the inputs of the digitally implemented combining circuits 340, where they are combined with the corresponding voltage modulation signals to generate a pair of modulation signals for each phase of the three-phase voltage. As shown in Figure 8, a first plurality of modulation signals 675 is provided from the circuits of the combiner 340 to the carrier reference circuit 345, and a second plurality of modulation signals 680 is provided from the circuits of the combiner 340 to the reference circuit. carrier 350. The outputs of the carrier reference circuit 345 are used as PWM control signals 355 to control the operation of the first inverter 120 (port activators 360 not shown). The outputs of the carrier reference circuit 350 are used as control signals from PWM 370 to control the operation of the second inverter 125 (port activators 360 not shown). EXAMPLIFIER CONTROL METHOD
[00124] Figure 9 shows a method 700 for controlling an energy conversion system. As shown, three-phase voltages are measured at 705 and provided with an optional low-pass filter at 707. The analog output of the low-pass filter is converted into digital signals at 710, which are then subject to sequence decomposition at 713 A voltage control algorithm is performed, in the dq coordinate system, on the decomposed signals in 715. The outputs of the voltage control algorithm are transformed into abc coordinates in 717. The resulting abc signals are used to generate the 720 voltage modulation signals.
[00125] In parallel with the operations shown in 705 to 720, method 700 performs the operations that are related to the circulating currents that flow through the transformers between cells. In 723, the circulating currents are measured and are subject to an optional low-pass filter operation in 725. The filtered analog signals are converted to digital signals in 727. The digital values of the circulating currents are passed to the current sharing controllers in 730. Current sharing controllers perform a series of operations in 735. Among these, current sharing controllers apply a DC cut-off filter, a low-pass resonant filter, and a fundamental frequency resonant filter to generate the current sharing modulation signals. At 740, the voltage modulation signals of 720 and the current sharing modulation signals of 735 are used to generate the PWM control signals. The PWM control signals are provided to the door activator circuits, which provide switching voltages to the inverters used in the energy conversion system. EXAMPLE SIMULATIONS
[00126] Figures 10a - 10b are graphs of exemplifying signals associated with the voltages (Vinva1, Vinvb1, Vinvc1 and Vinva2, Vinvb2, Vinvc2) of an energy conversion system that does not implement the control scheme indicated above. In figure 10a, the phase current 750 corresponds to the current generated because of the voltage outputs Vinva1, Vinvb1 and Vinvc1 of the inverter 120. The phase current 755 corresponds to the current generated because of the voltage outputs Vinva2, Vinvb2 and Vinvc2 of the second inverter 125. The resulting circulating currents 760 through the transformers between corresponding cells 175, 180 and 185 have a low frequency component that varies slowly over time compared to the fundamental frequency. This results in a correspondingly large variation in flow 765 (Figure 10c) of the transformer cores between cells 175, 180 and 185, which subjects the transformer cores between cells to potential saturation and limits the ability to design transformers between cells to use high permeability core materials.
[00127] A similar analysis applies to currents associated with neutral voltage Vn signals, which are shown in figure 10b. More particularly, the phase current 770 corresponds to the current generated because of the voltage output Vinvn1 of the first inverter 120, whereas the phase current 775 corresponds to the current generated because of the voltage Vinvn2 of the second inverter 125. The resulting circulating current 780 through the fourth transformer between cells 190 has a low frequency component that varies slowly over time. This results in a correspondingly large variation in flow 785 (Figure 10c) of the fourth transformer between cells 190, which subjects the nucleus of the fourth transformer between cells 190 to potential saturation, limiting the use of high permeability core materials in the fourth transformer between cells 190.
[00128] Figures 11a - 11b are graphs of exemplary signals associated with the voltages (Vinva1, Vinvb1, Vinvc1 and Vinva2, Vinvb2, Vinvc2) of the energy conversion system 100 that has the control scheme indicated above. In figure 11a, the phase current 800 corresponds to the current generated because of the voltages Vinva1, Vinvb1 and Vinvc1 of the first inverter 120, whereas the phase current 805 corresponds to the current generated because of the voltages Vinva2, Vinvb2 and Vinvc2 of the second inverter 125. As shown, the low frequency component in Figures 10a - 10b does not show the resulting circulating current 810 through the corresponding transformers between cells 175, 180 and 185. As a result, there is relatively little low frequency variation in flow 815 (Figure 11c) of transformers between cells and they can be designed using high permeability core materials.
[00129] A similar analysis applies with regard to the neutral voltage Vn of the energy conversion system 100. In figure 11b, the phase current 820 corresponds to the current generated because of the Vinvn1 voltage of the first inverter 120, whereas the phase current 825 corresponds to the current generated because of the Vinvn2 voltage of the second inverter 125. As shown, the low frequency component in Figures 10b - 10c does not show the resulting circulating current 830 through the fourth transformer between cells 190. As a result from this, there is relatively little low frequency variation in flow 835 (Figure 11c) of the fourth transformer between cells 190 and it can be designed when using high permeability core materials. EXAMPLE APPLICATION
[00130] The modalities of the energy conversion system 100 can be used in a wide variety of applications. Figure 12 describes how the energy conversion system 100 is incorporated in the context of the example method 1000. Figure 13 describes how the energy conversion system 100 can be incorporated in a plane 1005. During pre-production, the example method 1000 may include the specification and design 1010 of the aircraft 1005 and the acquisition of material 1015. During production, component and subassembly manufacturing 1020 and system integration 1025 of the aircraft 1005 take place. Then, the aircraft 1005 can proceed through certification and delivery 1030 to be put into service 1035. While in use by a customer, the 1005 aircraft is scheduled for routine maintenance and 1040 service (which may also include modification, reconfiguration, renewal, and so on) onwards, from the energy conversion system 100).
[00131] Each of the operations of the 1000 example method can be performed or performed by a system integrator, by third parties and / or by an operator (for example, a customer). For the purposes of this description, a system integrator may include without limitation any number of aircraft manufacturers and subcontractors to the main system; third parties may include without limitation any number of vendors, subcontractors and suppliers; and an operator can be an airline, a leasing company, a military entity, a service organization, and so on.
[00132] As shown in Figure 13, the aircraft 1005 produced by the example method 1000 can include a fuselage 1043 with a plurality of top-level systems 1045 and an interior 1050. Examples of high-level systems 1045 include one or more of a propulsion system 1055, an electrical system 1060, a hydraulic system 1065 and an environmental system 1070. The electrical system 1060 may include one or more energy conversion systems 100 of the type shown. The energy conversion system 100 can supply power to many of the high-level systems or other systems on the airplane 1005. In addition, the energy conversion system 100 can be included as part of the object of the method in Figure 11. Although an example aerospace is shown, the principles described may apply to other industries, such as the automotive industry, the computer industry and the like. The apparatus and methods presented here can be employed during any one or more stages of the example method 1000. For example, the components or subassemblies that correspond to the 1010 production process can be manufactured or manufactured in a similar way to the components or subassemblies. produced while the 1005 aircraft is in service. In addition, one or more modalities of the apparatus, method modalities or a combination of them can be used during the stages of production, for example, when substantially dispatching the set of or reducing the cost of a 1005 plane. more device modalities, method modalities or combination thereof can be used while the 1005 aircraft is in service, for example, and without limitation for maintenance and service 1040.

权利要求:
Claims (15)
[0001]
1. Energy conversion system (100) configured to supply multiphase energy that has a phase voltage associated with each phase of the multiphase energy respectively, characterized by the fact that the energy conversion system (100) comprises: a plurality of inverters (120, 125), comprising a first (120) and a second (125) inverters, configured to generate PWM output voltages for each phase voltage in response to PWM control signals (355, 370); a plurality of transformers between cells (660) configured to receive the output voltages PWM to generate the phase voltages, wherein the output voltages PWM result in a plurality of current flows in the plurality of transformers between cells (660), wherein the circulating current is a different current between the two output currents of each transformer between cells; a voltage controller (410) responsive to generate voltage modulation signals (675, 680) for each phase voltage; a plurality of current-sharing channels (330) associated respectively with each of the plurality of transformers between cells (660), wherein the plurality of current-sharing channels (330) is configured to generate current-sharing modulation signals (450, 460) in response to the plurality of circulating current flows; and a carrier reference circuit (345, 350) configured to generate the PWM control signals (355, 370) in response to the modulation signals, in which the modulation signals are obtained by combining the sharing modulation signals current (450, 460) and voltage modulation signals (675, 680), in which the plurality of transformers between cells (175, 180, 185, 190) is configured to supply three-phase supply voltages (605) when using voltages three-phase PWM outputs (605) interspersed between the first (120) and the second (125) inverters.
[0002]
2. Energy conversion system (100), according to claim 1, characterized by the fact that the first (120) and the second (125) inverters generate interchanged three-phase PWM output voltages.
[0003]
3. Energy conversion system (100), according to claim 1, characterized by the fact that each of the first (120) and the second (125) inverters additionally provides a neutral phase voltage.
[0004]
Energy conversion system (100) according to any one of claims 1 to 3, characterized in that it additionally comprises a current detection circuit (195) which has a plurality of current sensors (200, 210 , 220, 230) configured to provide circulating current signals to the current sharing channels (330) for each component of the plurality of inductive components.
[0005]
Energy conversion system (100) according to any one of claims 1 to 4, characterized in that the current sharing channels (330) comprise a fundamental resonant frequency controller that has a central frequency close to a fundamental frequency of the phase voltages.
[0006]
Energy conversion system (100) according to any one of claims 1 to 5, characterized by the fact that the current sharing channels (300) have a frequency transformation response that generally corresponds to:
[0007]
Energy conversion system (100) according to any one of claims 1 to 6, characterized by the fact that it also comprises a DC power supply (127) configured to supply input energy to the plurality of inverters, in that the plurality of inverters share a common DC bus.
[0008]
8. Energy conversion system (100) according to any one of claims 1 to 7, characterized by the fact that the inductive components are associated with the respective phases of the three-phase voltage signals (605), in which the sharing channels current (330) generate current sharing modulation signals (450, 460) associated with each phase of the three-phase voltage signals (605) respectively when using the plurality of circulating currents (760, 780) of the plurality of inductive components; and further comprises a plurality of combining circuits (340), in which each combining circuit is associated with a phase of the three-phase voltage signals (605), and in which the combining circuits (340) combine the voltage modulation signals (675, 680) and the current sharing modulation signals (450, 460) associated with each phase of the three-phase voltage signals (605) respectively to generate the PWM control signals (355, 370).
[0009]
Energy conversion system (100) according to any one of claims 1 to 8, characterized by the fact that it additionally comprises: a PWM activation circuit; a plurality of power supply output terminals coupled to the outputs of the plurality of inductive components; a plurality of current sensors (200, 210, 220, 230) coupled to the plurality of inductive components; a voltage controller (410) coupled to the plurality of inductive components; a current sharing system (330) coupled to the plurality of current sensors (200, 210, 220, 230); a combining circuit (340) coupled to the outputs of the voltage control system (325) and to the outputs of the current sharing system (330); and an activation circuit coupled to the combiner circuit (340).
[0010]
10. Energy conversion system (100), according to claim 9, characterized by the fact that the plurality of current sensors (200, 210, 220, 230) are coupled to the inductive components to supply the signals corresponding to twice the circulating current (760, 780) that flows through each of the inductive components.
[0011]
11. Energy conversion system (100) according to claim 9 or 10, characterized by the fact that the current sharing system comprises a plurality of current sharing controllers (440,445) arranged in pairs, in which each pair of the plurality of current sharing controllers is coupled to a respective current sensor (200, 210, 220, 230).
[0012]
Power conversion system (100) according to claim 9, 10 or 11, characterized by the fact that each current sharing controller (440, 445) of each pair of current sharing controllers (440, 445 ) comprises a resonant controller that has a central frequency close to a fundamental frequency of voltages in the plurality of power supply terminals.
[0013]
13. Method for controlling a multiphase energy conversion system (100) configured to supply multiphase energy having a phase voltage respectively associated with each phase of the multiphase energy, characterized by the fact that the method comprises: the generation, by a plurality of inverters (120, 125), with PWM output voltages for each phase voltage in response to PWM control signals (355, 370); the reception, by a plurality of transformers between cells (660), of the PWM output voltages to generate the phase voltages, where the PWM output voltages result in a plurality of circulating current flows in the plurality of transformers between cells (660), in which the circulating current is a current difference between the two output currents of each transformer between cells; the generation of voltage modulation signals (675, 680) for each phase voltage; the generation of current sharing modulation signals (450, 460) in response to the plurality of circulating current flows; the generation of PWM control signals (355, 370) in response to the modulation signals, in which the modulation signals are obtained by combining the current sharing modulation signals (450, 460) and the voltage modulation signals (675, 680); and the supply, by the plurality of transformers between cells (175, 180, 185, 190), of three-phase supply voltages (605) using PWM output voltages (605) interspersed from the first (120) and second (125) inverters.
[0014]
14. Method, according to claim 13, characterized by the fact that the generation of current sharing modulation signals comprises the application of a resonant controller at a central frequency close to a fundamental frequency of each phase of the multiphase energy.
[0015]
15. Method, according to claim 14 or 15, characterized by the fact that the generation of current sharing modulation signals comprises the application of a controller that has a transformation frequency response that in general corresponds to: DC Low frequency Fundamental frequency where wi defines a central frequency of a low frequency filter, wf defines a central frequency of a fundamental frequency filter, Δwl defines a bandwidth of the low frequency filter, Δwf defines a bandwidth of the fundamental frequency filter , Kl0 and Kf0 define the pass band magnitudes of the low frequency filter and the fundamental frequency filter, respectively, Kl and Kf define peak gains of the low frequency filter and the fundamental frequency filter, respectively, and Cph (s) is a phase delay compensator that provides phase compensation around the fundamental frequency.

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同族专利:

公开号 | 公开日

CN103973145B|2018-09-25|

EP2768129A2|2014-08-20|

US20140211522A1|2014-07-31|

RU2014101396A|2015-07-27|

JP6422216B2|2018-11-14|

JP2014147283A|2014-08-14|

CA2836335C|2016-10-18|

EP2768129A3|2015-11-04|

CA2836335A1|2014-07-29|

US8964432B2|2015-02-24|

EP2768129B1|2019-02-20|

RU2620582C2|2017-05-29|

CN103973145A|2014-08-06|

BR102014001743A2|2015-07-28|

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法律状态:
2015-07-28| B03A| Publication of a patent application or of a certificate of addition of invention [chapter 3.1 patent gazette]|

2018-11-13| B06F| Objections, documents and/or translations needed after an examination request according [chapter 6.6 patent gazette]|

2020-01-28| B06U| Preliminary requirement: requests with searches performed by other patent offices: procedure suspended [chapter 6.21 patent gazette]|

2020-12-15| B09A| Decision: intention to grant|

2021-02-09| B16A| Patent or certificate of addition of invention granted|Free format text: PRAZO DE VALIDADE: 20 (VINTE) ANOS CONTADOS A PARTIR DE 24/01/2014, OBSERVADAS AS CONDICOES LEGAIS. |

优先权:

申请号 | 申请日 | 专利标题

US13/752,813|US8964432B2|2013-01-29|2013-01-29|Apparatus and method for controlling circulating current in an inverter system|

US13/752,813|2013-01-29|

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